Geographic locating remote endpoint monitor device, system, and methodology thereof

ABSTRACT

A phase-locked loop frequency synthesizer includes an L-state pulse width modulator configured to receive a reference frequency signal and at least one entry from a frequency table, and to output at least one N/N+1 modulus signals corresponding to the at least one entry from the frequency table. The synthesizer includes a divide by N/N+1 controllable modulus divider configured to receive the at least one N/N+1 modulus signals and to divide the output frequency signal by the at least one N/N+1 modulus signals to generate a second reference frequency signal. The synthesizer includes a phase frequency detector configured to receive the reference frequency signal and the second reference frequency signal and to generate an error signal. The synthesizer also includes a filter network configured to receive the error signal and to output a voltage; and a voltage controlled oscillator configured to receive the voltage and to generate the output frequency signal.

CROSS REFERENCE TO RELATED PATENT DOCUMENTS

The present application is a divisional of application Ser. No.13/740,806 filed Jan. 14, 2013, which is a continuation of applicationSer. No. 12/340,272 filed Dec. 19, 2008, now U.S. Pat. No. 8,374,228issued Feb. 12, 2013, which is a continuation of application Ser. No.10/662,530 filed Sep. 16, 2003, now U.S. Pat. No. 7,477,694 issued Jan.13, 2009, which is a continuation of and is based upon and claimspriority from PCT application No. PCT/US02/06674 filed Mar. 29, 2002,all of which are incorporated herein by reference. The present documentclaims the benefit of the earlier filing date of U.S. provisional patentapplication Ser. No. 60/279,671, entitled “ENHANCED WIRELESS PACKET DATACOMMUNICATION SYSTEM, METHOD, AND APPARATUS APPLICABLE TO BOTH WIDE AREANETWORKS AND LOCAL AREA NETWORKS,” filed in the United States Patent andTrademark Office on Mar. 30, 2001, and having common inventors as thepresent document, the entire contents of which is incorporated herein byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention is directed toward communication related systems,wide area and local area networks, apparatuses and methods as well ascomputer based digital signal processing mechanisms and methods usedtherein. More particularly, the invention is directed towardcommunication systems, methods, and apparatuses that use signals thatare modulated using a combination of FSK modulation and QAMsubmodulation.

2. Discussion of the Background:

It would be advantageous if a remote environmental monitor, a utilitymeter (e.g., water, gas, or electric), a security system, a mobile dataapplication, or other remote device, hereafter referred to as a “remoteendpoint,” could communicate with a base station receiver which in turnwould forward the endpoint information (e.g., impurity count, meterread, alarm, or position fix) to some central location. This centrallocation could be part of a wide area network (WAN), or a local areanetwork (LAN).

WAN Applications Cost to Acquire Remote Endpoint Data

The cost to acquire information from remote endpoints includes (using aremote meter monitoring application as an example) the cost of the meterinterface, the meter transmitter (or transceiver), the remote receiver(or transceiver), as well as the forwarding infrastructure, includingradio spectrum costs. For example, if the remote receiver has a verylimited range such that it can only communicate with one meter, then thecost of that receiver must be fully burdened into the cost to acquirethe meter information. Additionally, the cost to install and maintainthat remote receiver must be fully burdened as well.

It would therefore be desirable to have a remote receiver (transceiver)communicate with as many endpoints as possible. It would be advantageousif the remote receivers were each able to cover as great a range aspossible. Having sufficient range would eliminate costly intermediatetiers of repeaters and/or store-and-forward hand-off devices. It wouldbe advantageous if a radio system that could communicate over many mileswere very reliable since the loss of a base station that is responsiblefor receiving data from thousands or tens of thousands of remoteendpoints could be devastating.

Furthermore, it would advantageous if the system were not readilysusceptible to jamming. Any source of interference within “view” (i.e.,listening range) of a receiver can severely degrade the reliability ofthe system. Such degradation in reliability translates into a reducedrange of the system. A five-block range creates a view area ofapproximately 0.8 square miles. A 10-mile range creates a view of 314square miles. Accordingly, an increase in range brings a quadraticincrease in the probability of an in-view interference source (the areain a circle is a function of the square of the range).

Conventional approaches for wide area remote monitoring, may be broadlycategorized into three groups: short range, long range, and very longrange (i.e., satellite).

Short Range Products (One Mile or Less)

Products falling into this category generally have high infrastructurecosts associated with them. Large numbers of remote receivers must bebought, installed and maintained. By their nature, short-range devicesrequire either a receiver dedicated to a single endpoint, orintermediate transceivers set up in a cellular or repeater fashion tocreate multiple tiers of data handoff. Some shorter-range strategiesrequire that a receiver be co-located with a cable box, home computer,modem, or in a drive-by vehicle. These approaches do not yieldubiquitous coverage area and/or are poorly controlled and thus not areliable alternative.

Long Range Products (One Mile or More)

Conventional long-range communications services for multi-point-to-pointmonitoring are based on two-way communications. Two-way handshaking isneeded to dynamically assign frequency channel usage amongst customers.In the case of voice and two-way messaging, it is a logical requirement(users “talk” and “listen”)

Very Long Range Products (satellite)

A satellite's greatest advantage, range, is also a weakness. As therange (footprint) increases, the number of endpoints in view increasesquadratically, and so does required bandwidth. A competitive satellitesystem would require approximately 200 MHz of bandwidth to equate to 500KHz of terrestrial base stations. This will considerably impedesatellites from servicing dense applications that require updates every15 minutes. In addition, present satellite systems are two-way (arequirement for licensed frequency dynamic allocation). This creates apermanent three times or greater cost disadvantage for such systems(present satellite transceivers have at least a 20 times costdisadvantage due to other factors such as very tight ppm budgets).Further, the required update rates would yield unacceptable batterylife.

Most of the LAN data communications suppliers and standards (IEEE802.11, Bluetooth, Swap Home RF, ultrawide bandwidth, etc.) are focusedon the high data rate (1-10 million bits per second) market required forPC-to-PC connection.

These higher data rates significantly reduce range and therefore requiremore closely spaced base stations. This in turn increases hardware andinstallation cost. This also can increase the requirement for repeaters.

SUMMARY OF THE INVENTION

The inventors of the present invention have recognized that conventionalapproaches to remote monitoring are inadequate from both a technical anda cost effectiveness perspective. Accordingly, an object of the presentinvention is to address these identified inadequacies, as well asothers, to provide a system for remote monitoring that has advantageousperformance characteristics, is reliable, and is cost effective makingit an option for a vast range of potential applications for remotemonitoring.

The inventors of the present invention have recognized that remotemonitoring solutions should not be bound to either a WAN or a LANenvironment. Accordingly, the present inventors have devised an approachto remote monitoring that is not limited to any network topology, butrather, may be customized to meet the unique needs of any installation.The present invention is applicable to systems that exist in a LANenvironment, or a WAN environment. A WAN in the present invention can beany large geographic area, for example an entire metropolitan area. ALAN in the instant invention may be any smaller environment, for examplea commercial building, an industrial plant or a campus of contiguousbuildings and facilities.

The inventors of the present invention have further recognized that theuse of middle layer repeaters limits the flexibility of a remotemonitoring system, and for some applications, makes the remotemonitoring cost-prohibitive. Accordingly, the present invention does notmandate a middle-tier repeater layer and enjoys a significantly lowercost infrastructure than the existing alternatives.

The inventors of the present invention have further recognized thatone-way communication systems may be built at a significant cost savingsand complexity savings over bi-directional communication systems.Accordingly, the present invention primarily uses one-way transmit-onlydevices with the advantage of three times or greater product costsavings over a two-way device. Furthermore, the present inventors haverecognized that a transmit-only architecture provides a huge batterylife advantage over two-way systems that must also power a receiver.This advantage makes the present invention an attractive option forcertain applications that heretofore have been unable to practicallyconsider remote monitoring. Further yet, two-way systems typically havea longer synchronization/handshaking time. This translates into longeron-air time, use of greater bandwidth and shorter battery life.

One embodiment of the present invention operates at 16.6 kb/s and iswell suited to ranges between 5 to 100 kb/s. This is advantageous fromseveral perspectives. First, higher data rates translate into lowerrange. Systems operating at a 2 megabit per second data rate would havereduced range corresponding to the instant invention transmitting atabout 100 times less power (10*log(2×10⁶/16.6×10³)). Second, lower datarate products that are power pole mounted, such as Whisper, must remainon the air longer than the devices of the present invention(approximately 20 times or more, due to data rate and initial preambleacquisition), and are not optimized to operate in a transmit-only mode.This reduces the number of remote endpoints that can be read by onereceiver, increases infrastructure cost, and considerably reducesbattery life in applications such as water and gas meters. Higher datarate (e.g., 128 kb/s), power pole mounted products, such as Metrocom'sRicochet, have other deficiencies. Ricochet systems must repeat datamessages from pole mounted data collectors in a bucket brigade fashion.This wastes radio bandwidth and adds complex routing hardware andsoftware.

One WAN-based embodiment of the present invention is configured tooperate in licensed frequency bands such as 218 MHz, 220 MHz and 700MHz.

The present invention is equally applicable to local area networkenvironments that can have simple point-to-point, or more complexmulti-point-to-point configurations. The ‘point’ can be a small basestation or a network of interconnecting base stations. Thoseinterconnections can be hard wired or wireless where the base stationsform interconnections via smart and dumb repeaters. One LAN-basedembodiment of the present invention is configured to operate primarilyin unlicensed bands such as 868 MHz, 915 MHz, 2.4 GHz and 5.8 GHz.

According to the present invention, several novel methods are combinedto reduce remote endpoint cost, and to increase the system's overallrange (e.g., up to 10 miles, or even greater distances). This increasedrange in turn leads to the feasibility of a novel system architecturewhich includes geographically dispersed tower receiver base stationswhich connect to one or more data concentrators.

In one embodiment of the present invention, range is increased by usinga novel form of FSK data modulation that eliminates the effect ofmessage DC, allowing direct VCO modulation at full data rate. Theinventors of the present invention have also invented novel DSP receiveralgorithms that achieve near theoretic sensitivity at full data ratewhile rapidly eliminating the frequency uncertainty caused by low costcrystals used in the remote endpoint.

Conventional systems typically achieve much shorter ranges and,therefore, quadratically smaller service coverage areas. This shorterrange in operation corresponds to a quadratically smaller number ofmonitorable endpoint devices. It also corresponds to a quadraticallylarger number of base station receivers required to cover a region, andan increase in the infrastructure required for data collection. Forexample, in Arthur et al. (U.S. Pat. No. 4,977,577), Sanderford et al.,(U.S. Pat. No. 5,987,058), Rouquette et al. (U.S. Pat. No. 5,920,589),and Sanderford et al., (U.S. Pat. No. 5,953,368) a range ofapproximately 1/2 mile was achieved when monitoring a data collectorfrom a power pole approximately 30 feet off of the ground. The resultingcoverage area was 0.79 square miles. The present invention has anincreased range of at least 10 miles when an antenna is elevated 200feet on an existing communications tower. The resulting coverage areawith the present invention is 314 square miles, or 400 times thecoverage area of conventional systems. The inventors of the presentinvention have recognized that by achieving such an expansive range,that it is economically practical to provide 3× or more overlappingcoverage, which provides excellent signal path redundancy. It wouldrequire 400 data collectors based on conventional systems with noredundancy overlap to match the coverage area of one data collectorbased on the present invention. Adding overlap to achieve theperformance of the present invention only increases this number. Inorder to cover a large city using conventional systems, thousands ofpower pole mounted data collectors would be needed. A further deficiencyof conventional data collectors is that they must be connected by somemethod, but it is cost prohibitive to use copper or fiber connection.

It is impractical to install, pay for and maintain thousands oflandline/phone connections in order to collect data at a central point.Conventional systems use RF repeater or other RF backhaul means. Thiswastes bandwidth, can double hardware costs (can require a second set ofradio hardware) at each power pole mounted data collector and typicallyleads to complex routers and/or smart databases to reduce communicationstraffic on the backhaul.

The inventors of the present invention have further recognized that byachieving such gains in coverage range, so few tower base stations arerequired that system architectures can be configured that are veryelegant yet completely scalable. In one embodiment of the presentinvention, each tower mounted base station uses a landline connecteddirectly to a centralized data concentrator. A typical city may becovered with 10 to 14 tower base stations, a city the size of Dallaswith 40 to 50.

The inventors of the present invention have recognized that theproliferation of communication towers due to the expansion of thecellular telephone industry provides an advantageous alternative to polemounting. Ten years ago, the construction of a tower and right-of-wayacquisition may cost $125,000 to $250,000 each. Today, space on acommunication tower can be rented for $500 to $1,000/month. In addition,these towers are already in ideal locations, elevations and ergonomicdeployments required to meet the need of cellular and othercommunications systems. Further, these tower sites have existing powerand phone lines accessible, and do not require the permission of localutilities to access.

The present invention achieves excellent signal path redundancy whilegreatly simplifying the message routing and hand-off overhead typical inconventional cellular phone systems. Cellular telephone systems must betwo-way in order to provide two-way voice communication, and frequencychannel allocation. The present invention, in some embodiments, isimplemented as a one-way communication system, and in other embodimentsas a half duplex two-way. Conventional cellular telephone systemscarefully design terrain coverage to minimize the overlap of one cellinto another. Furthermore, conventional cellular telephone systems use 7frequency channels to insure that each adjacent cell operates on adifferent frequency than its neighbors. This method is well known in theart and referred to as frequency re-use. The inventors of the presentinvention have recognized that by reusing the same frequencies inadjacent cells, that a 7× efficiency of frequency usage can be achievedin an ALOHA system.

Cellular phone systems therefore, do not typically enjoy the benefits ofsimultaneous redundant reception of a remote phone conversation.However, as recognized by the present inventors, this redundancy wouldbe very valuable in a data system (especially a transmit-only one) wherethere is no caller to say, “can you repeat that last sentence.” As anexample, if a single channel were 95% reliable in a conventional system,the effect of a 3× tower base station coverage according to the systemof the present invention would be given by:

P(success)/conventional=0.95

P(failure)_(conventional)=1−0.95=0.05 (single base station)

P(failure)_(present invention)=0.05×0.05×0.05=0.000125

P(success)_(present invention)=1−000125=0.999875

In addition to drastically reduced infrastructure cost and increasedreliability, the instant invention architecture provides resistance tosignal fading, shading, vandalism, near-far effect (power control),mobile hand-off, and more as described herein.

The present invention relates to a communication system using techniquesand devices that are flexible in scope, design, and capability. Further,the present invention relates to communication systems that may beconfigured for various applications. For example, the present inventionrelates to communication systems that are configured for point-to-pointone-way communications (e.g., transmitter to receiver), point-to-pointtwo-way communications (e.g., transceiver to transceiver), many-to-oneone-way communications (e.g., transmitter to receiver), many-to-onetwo-way communications (e.g., transceiver to transceiver), point-to-manyone-way communications (e.g., many receiving endpoints, including eitheroverlapping or non-overlapping receive areas of coverage), point-to-manytwo-way communications (e.g., transceiver to transceiver, includingeither overlapping or non-overlapping coverage areas) many-to-manyone-way communications (e.g., transmitter to receiver, including eitheroverlapping or non-overlapping receive areas of coverage), or repeaterapplications including any combination of transmitter, receiver andtransceiver devices.

In one embodiment, the present invention relates to cost-effectivetransmitters and receiver architectures that use readily availablecomponents to implement novel narrowband modulation and demodulationtechniques to provide a superior data communication capability. Bursttelemetry is efficiently transmitted and received using packet dataradio communications. The data is communicated in a bandwidth efficientmanner maximizing a data rate per unit of bandwidth. Themodulation/demodulation techniques described herein are designed toprovide an extended communication range and to avoid a false signaldetection (also referred to herein as a false ‘trip’).

Both frequency domain and time domain modulation techniques are used tocreate a modulation domain signal that can be used at the receiver tominimize false trips. The modulation techniques of the present inventionprovide for improvements over traditional narrowband FSK, M-ary FSK,BPSK, QPSK, or other constant-envelope techniques for transmitting andreceiving burst data as packet data radio signals.

The present invention uses angle modulation (i.e., constant-envelope)techniques, which can be accomplished using low cost, nonlinearamplification techniques at the transmitter. The present invention usesa modulation technique that combines M-ary FSK with a QAM submodulation,both of which may be implemented with low cost components at thetransmitter.

The present invention allows for a method of shaping the transmitspectrum in a constant envelope method that suppresses spectral regrowtheven if nonlinear amplification methods are employed. The resultingbenefit is that typical class-C type amplification techniques can beused for transmitting the modulated signal without losing the benefit ofthe data rate to spectral use ratio. This fundamentally provides costand power utilization benefit to the transmitter.

The present invention provides for a method of efficiently maximizingthe data rate in a given bandwidth of frequency utilization whileproviding excellent signal detection and false trip avoidance at thereceive component.

The present invention compensates for frequency error between thetransmitter and receiver components with a minimum of link budgetimpact. The frequency error tolerant system described herein providesfor low cost system frequency references at transmit and receiveendpoints. Methods described herein define a way for the system toachieve maximum sensitivity and corresponding range while simultaneouslyallowing for frequency error without the need for automatic frequencycontrol (AFC) or Costas loop (feedback) frequency correction. Thepresent invention in one embodiment employs processing methods at thereceive component which are feed-forward, achieving sensitivityapproaching closed loop coherent systems without incurring the cost orthe long acquisition times of the feedback elements. Alternatively, butwith added cost and energy consumption, the frequency error can beeliminated using classical feedback error correcting methods thussimplifying the FM demodulator.

The present invention provides for methods of resolving narrow frequencyerrors as well as wide frequency errors. Narrow frequency errors arethose errors that occur inside the passband of the received processingthread while wide frequency errors force the receiver component to adaptalternative techniques described herein.

The present invention provides for single channel as well asmultichannel receive architectures, detailing cost effective methods forimplementing wide-area cellular as well as point-to-point networks.

The present invention provides for low power transmit components andmethods to conserve battery life and thus reduce transmitter cost andpower requirements.

The present invention provides dynamic and adaptive signal detection andacquisition techniques which utilize parameters of the transmitmodulation to achieve superior false trip avoidance and rapid signalacquisition.

The present invention provides for transmit-only endpoints without powercontrol due to the wide dynamic range receive component which receivespacket data, burst messages. The transmitters are fundamentally low costsince they do not require a return communication channel nor powercontrol.

The present invention has utility in point-to-point applications as wellas cellular and redundant cellular topologies. All applicationtopologies enjoy a fundamentally low cost implementation that can usethe transmit-only endpoint devices without power control.

The present invention describes a method for producing the transmitcarrier frequency using extremely low cost pulse width modulation (PWM)components readily available as a sub-element in many commerciallyavailable microcontrollers. The methods described herein provide for lowcost transmitters using commonly available elements. Furthermore,additional methods described herein detail ways to compensate at thereceive component for signal degradation created in the fundamentallylow cost transmitter due to non-ideal elements.

Consistent with the title of this section, the above summary is notintended to be an exhaustive discussion of all the features orembodiments of the present invention. A more complete, although notnecessarily exhaustive, description of the features and embodiments ofthe invention is found in the section entitled “DESCRIPTION OF THEPREFERRED EMBODIMENTS.”

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete appreciation of the present invention and many of theattendant advantages thereof will be readily obtained as the samebecomes better understood by reference to the following detaileddescription when considered in connection with the accompanyingdrawings, wherein:

FIG. 1 illustrates base stations using overlapped coverage areasconnected to a common data concentrator;

FIG. 2 illustrates colliding data packets;

FIG. 3 illustrates near-far effects, which benefit co-channelperformance;

FIG. 4 illustrates frequency channel usage load leveling;

FIG. 5 illustrates bandwidth efficient, non-channelized spectrumutilization;

FIG. 6 illustrates conventional cellular system area coverage;

FIG. 7 illustrates a cellular layout according to one embodiment of thepresent invention;

FIG. 8 illustrates a message format;

FIG. 9 illustrates differential history values contained in datamessage;

FIG. 10 illustrates single bit error correction;

FIG. 11 illustrates message sequence numbers for missed message;

FIG. 12 illustrates data concentrator operation;

FIG. 13 illustrates conventional cellular radio system using sectoredantennas to increase capacity;

FIG. 14 illustrates omni-directional antenna usage;

FIG. 15 illustrates base station configuration;

FIG. 16 illustrates a transmitter with alternate configurations;

FIG. 17 illustrates a method to allow wide geographic freedom overdifferent licensed frequency bands in a mobile tracking system withoutthe need of externally controlled frequency switchover and management;

FIG. 18 illustrates an example of automatic frequency selection for atransmit-only system sending GPS data for remote location;

FIG. 19 illustrates smart repeater usage;

FIG. 20 illustrates selectable enhanced signal margin without the costof higher output power transmitter amplifier stages;

FIG. 21 illustrates additional area covered by boost mode;

FIG. 22 illustrates boost mode channel load leveling;

FIG. 23 illustrates assigning separate boost channel to avoid thechannel roll-off of strong transmitter in adjacent channels;

FIG. 24 illustrates signal bandwidth vs. guard band;

FIG. 25 illustrates signal bandwidth vs. guard band in “boost” mode;

FIG. 26 illustrates transmitter locations that require boost mode;

FIG. 27 illustrates a 10-channel frequency plan;

FIG. 28 illustrates transmission duty cycle used to calculate batterylife;

FIG. 29 is a block diagram of a basic base station system configuration;

FIG. 30 illustrates remote processor program transfer protocol message;

FIG. 31 illustrates LAN to WAN bridge;

FIG. 32 illustrates geographic boundary of WAN radio license;

FIG. 33 illustrates LAN to WAN bridge example;

FIG. 34 illustrates special exception LAN messages transferred to WAN;

FIG. 35 illustrates remote serial data monitoring;

FIG. 36 illustrates a basic FSK transmitter block diagram;

FIG. 37 illustrates 16QAM submodulator constellation andfrequency-domain representation;

FIG. 38 illustrates a fractional-N phase locked frequency synthesizerwith PWM divider modulus control;

FIG. 39 illustrates basic r theta trip algorithm;

FIG. 40 illustrates additional delay elements for r theta tripalgorithm;

FIG. 41 illustrates a “dual mode” QAM correlator;

FIG. 42 illustrates coefficients for a “dual mode” QAM correlator;

FIG. 43 illustrates narrow r acceptance test;

FIG. 44 illustrates wide r acceptance test;

FIG. 45 illustrates theta/angle acceptance test;

FIG. 46 illustrates r theta trip determination;

FIG. 47 is a block diagram of a system for signal acquisition anddemodulation;

FIG. 48 illustrates data decision threshold scale factor;

FIG. 49 illustrates adaptive slicer algorithm;

FIG. 50 illustrates a wide error frequency resolving trip algorithm;

FIG. 51 is a block diagram illustrating a wide error frequency resolvingreceiver;

FIG. 52 illustrates a 7FSK/16QAM receiver with a “dual-mode” correlator;

FIG. 53 is a multi-frequency pipelining diagram;

FIG. 54 illustrates a demodulation equalizer option for the system ofFIG. 52;

FIG. 55 illustrates a receiver architecture using a single RF front-end,single A/D and multiple channel demodulation;

FIG. 56 illustrates a receiver architecture including a single RFfront-end with multiple A/D and channel demodulation;

FIG. 57 illustrates a basic receiver block diagram;

FIG. 58 illustrates channel filter options;

FIG. 59 illustrates the receiver of FIG. 57 with image rejecting analogdownconversion;

FIG. 60 illustrates a receiver using multiple downconversions;

FIG. 61 illustrates a five-tap differentiator;

FIG. 62 illustrates a seventeen-tap differentiator;

FIG. 63 illustrates 7FSK/16QAM symbol patterns;

FIG. 64 is a block diagram of a basic PSK transmitter according to oneembodiment of the present invention;

FIG. 65 is a block diagram of a basic PSK receiver according to oneembodiment of the present invention;

FIG. 66 is a block diagram of a basic ASK transmitter according to oneembodiment of the present invention;

FIG. 67 is a block diagram of a basic ASK receiver according to oneembodiment of the present invention; and

FIG. 68 is an exemplary computer system programmed to perform one ormore of the special purpose functions of the present invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

In designing conventional cellular networks, steps are taken to ensurethat adjacent cells (receivers) do not compete for a given transmitteron the same frequency. In the case of a conventional direct sequencespread spectrum communication system using CDMA, the receive devicesmust power control the transmit components in order to accommodatemultiple users.

In one embodiment of the present invention, endpoint devices aretransmit-only, and do not require power control by a receiver. Moreover,adjacent cell towers may each monitor a single endpoint at the same timeusing the same frequency.

FIG. 1 illustrates a system according to one embodiment of the presentinvention. As shown in FIG. 1, the system includes several cellular basestations HA10, HA20, HA30 which are geographically dispersed. In oneembodiment of the present invention, the base stations HA10, HA20, HA30are dispersed across a metropolitan area. The system also includes oneor more endpoints HA40, HA50 that include a transmitter. In oneembodiment of the present invention, the endpoints HA40, HA50 aretransmit-only devices. In an alternative embodiment, the endpoints HA40,HA50 are transceivers, which permits bi-directional communication. Theendpoints HA40, HA50 are within range of at least one, and possiblymultiple base stations HA10, HA20, HA30. The base stations HA10, HA20,HA30 transmit each message they receive to a Data Concentrator HA60.Messages are transmitted to the Data Concentrator HA60 through a directserial connection in one embodiment, although other approaches 10 couldbe used including, but not limited to, single phone lines (POTS/PPP),dual phone lines (Dual POTS/multilink PPP), Ethernet, ISDN, a microwaveor other wireless RF link, and leased lines (frame relay).

The Data Concentrator HA60 receives all transmitted messages from allendpoints HA40, HA50 within range of all base stations HA10, HA20, HA30connected to the Data Concentrator HA60. Since the Data ConcentratorHA60 receives all messages received from all base stations HA10, HA20,HA30, and more than one base station HA 10, HA20, HA30 may receive anygiven message from an endpoint HA40, HA50, the Data Concentrator HA60often receives redundant messages. The Data Concentrator eliminatesthose redundant messages. By adding an identification parameter to eachmessage corresponding to the endpoint HA40, HA50 transmitting themessage, the elimination of redundant messages in the Data ConcentratorHA60 is facilitated. The identification parameters may include, forexample, data identification tags in the payload, or overhead bits.

In one embodiment of the present invention, multiple base stations HA10,HA20, HA30 operate on the same frequency, regardless of whether thecoverage regions of the base stations HA10, HA20, HA30 overlap. Byallowing the coverage regions of the base stations HA10, HA20, HA30 tooverlap, reliability is improved as each transmitting endpoint HA40,HA50 may be observed by more than one base station HA10, HA20, HA30. Ifa particular base station HAL0, HA20, HA30 becomes unusable due tosignal degradation caused by interference, collision or obstruction, itis likely that an alternative base station HA10, HA20, HA30 will succeedin collecting data for those endpoints.

In one embodiment of the present invention, the location of thetransmitting endpoint HA40, HA50 is roughly determined by the basestation HA10, HA20, HA30 receiver by correlating a transmitter IDincluded in a received message with a received signal strength usingtriangulation techniques understood by those of ordinary skill in theart.

In another embodiment of the present invention, the endpoints HA40, HA50include transceivers (i.e., for bi-directional communication). In thisembodiment, the received signal strength or history of throughputcollected at the various base stations HA10, HA20, HA30 is used todetermine which of the base stations HA10, HA20, HA30 is to be used tocommunicate back to a particular endpoint HA40, HA50 device. In otherembodiments, alternative approaches for two-way communication are used,including, for example, a positive handshake.

One advantage of the present invention, as shown in FIG. 1, is that itworks well for transmit-only devices as the endpoints HA40, HA50 withoutthe need for power control. As would be understood, two co-located,perfectly synchronous transmitters operating on the same frequency willeffectively jam each other with respect to any given receiver. By timestaggering the transmitters using pseudo-random means, the system of thepresent invention allows each transmitter endpoint HA40, HA50 theopportunity to communicate with a probability of success that is afunction of the duty cycle of the transmitter. In the case where twoendpoints HA40, HA50 transmit at the exact same instance, the fact thatthe endpoints HA40, HA50 are geographically separated makes it unlikelythat any receiver will receive two transmissions at the same signalstrength. Accordingly, greater system throughput is achieved since thestronger of the simultaneous receptions will be received by the basestation stations HA10, HA20, HA30 due to FM capture or other SNRadvantage. By adding more base stations HA10, HA20, HA30 (i.e.,receivers), even greater system throughput is achieved.

As discussed above, having overlapping cell sites monitoring the samefrequency increases system throughput. In other embodiments of thepresent invention, system throughput is increased through a data repeatscheme. By configuring the endpoints HA40, HA50 to redundantly transmiteach message separated by a pseudo random amount of time, theprobability of a successful transmission increases, and throughput isincreased.

In one embodiment of the present invention, the system is a many-to-onesystem as depicted in HA, where the endpoint HA40, HA50 transmittershave a low duty cycle, thereby allowing for many transmitters to share agiven frequency without a degradation of system throughput. Otheradvantages are also derived from having a low duty cycle, including, butnot limited to, the ability for each endpoint HA40, HA50 device to bebattery operated since the device is in a reduced power state for themajority its service life. If the duty cycle of the endpoints HA40, HA50is very low (e.g., less than 1%, and preferably much smaller), theeffective number of endpoints HA40, HA50 can become much larger whilestill allowing for the use of low-cost system timing components. In oneembodiment of the present invention, the lowest possible cost approachis used. One way of keeping cost down is by using a pseudo-randomtransmission interval on the transmit-only endpoint HA40, HA50 whichdoes not require any time synchronization as would be required byconventional time division multiple access (TDMA) cellular systems.

By using low duty cycle transmit devices for the endpoints HA40, HA50,power control is not required, since each base station HA10, HA20, HA30receiver need only listen to one transmitter at a time on a givenfrequency. The receivers of the present invention are designed tosimultaneously pass strong and weak signals without distortion.Reception of a weaker signal is aborted in order to acquire a strongersignal in the event of collision. Accordingly, the inventive systemtakes advantage of having base stations HA10, HA20, HA30 that aregeographically spaced, yet overlapping in coverage to resolvetraditional near-far problems associated with the simultaneous receptionfrom both a strong and weak transmitter. As shown in FIG. 1, transmitterHA50 is closer to base station HA20 relative to transmitter HA40. Inthis case, the base station HA20 is more likely to receive atransmission from HA50 even if the transmitter HA40 was operatingsimultaneously on the same frequency. By geographically dispersing thebase stations HAL0, HA20, HA30, a transmission that was aborted by HA20,for example, would be received by base station HA10 and/or base stationHA30.

Receiver Collision Processing

In one embodiment of the present invention, the abort algorithm,described above in the context of FIG. 1, is implemented in thereceiver. The abort algorithm uses an indicator of received signalstrength in making abort decisions. FIG. 2 illustrates the abortalgorithm in terms of two exemplary cases, Case 1, and Case 2, whichillustrate two of the most common collision events.

As shown in FIG. 2, Case 1 illustrates a case where a single receiverreceives a first weak signal HL110, followed by a second, strongersignal HL120. The second, stronger signal HL120 is received prior to thecompletion of the first, weaker signal HL110. The abort algorithmaccording to one embodiment of the present invention acts to abort thereception of the first, weaker signal HL110, and immediately beginprocessing the later, stronger signal HL120, making this determinationbased on received signal strength.

In other embodiments, other parameters are used in determining that thefirst signal HL110 being received is no longer worth decoding. Thereceiver architectures according to the present invention processreceived signals in the modulation domain whereby a mean frequency errorof a given transmitter may be measured. Should a second signal having adifferent receive strength be received, it is likely that the new signalwill have a different mean frequency error since it originated from adifferent imprecise frequency source. The different mean frequency errormay be used to re-start the data demodulation process. As shown in Case1 of FIG. 2, the two received signal can be viewed in terms of eithersignal strength or mean frequency error.

Case 2 of FIG. 2 illustrates the case where a collision occurs when aweaker signal HL150 is received after initial trip and acquisition of astronger signal HL140. Signal processing will typically prevent theobservation of the weaker signal HL150 until the stronger signal HL140completes its message (i.e., at the time indicated by HL160). In thissituation, the weaker message HL150 is typically lost unless the overlapis small and a leader portion of the second signal HL150 remains longenough for acquisition of that signal to occur.

The above-described combined abort algorithm of the present inventionprevents the loss of data from both received packets, and therefore,increases a probability of packet success from a probability following anon-slotted ALOHA form:

Non-Slotted ALOHA: Ps=1−[1−e ^(−2λNT)]^(M)

to a greater probability following a slotted ALOHA form:

Slotted ALOHA: Ps=1−[1−e ^(−λNT)]^(M)

where the term definitions are defined as:

Ps=probability of packet success;

=1/average time interval between transmissions;

N=number of simultaneous transmissions−1;

T=duration of single packet; and

M=redundancy of a packet at endpoint.

Non-slotted ALOHA and slotted ALOHA schemes are understood by those ofordinary skill in the digital communications art, and are described indetail in Sklar, B., “Digital Communications: Fundamentals andApplications,” Second Edition, ISBN 0-13-084788-7, Prentice Hall, 2001,the entire contents of which is incorporated herein by reference.

Accordingly, overall receiver throughput is improved dramatically withthe addition of an abort capability. By adding additional receiversand/or operational frequencies, exponential terms of the above equationsare increased, thereby even further improving throughput.

Typically, data message lengths in packetized data applications are in arange of about 10 to 30 milliseconds (msec) with messages occurringevery few seconds, minutes, hours or longer. The present inventionoperates with a standard data rate of 16.6 kbps. Even with messagesoccurring at a 30 second interval, the transmitter duty cycle remainslow (0.0006666). System loading per application dictates an acceptableduty cycle based on the number of transmitters, number of receivers, andnumber of available operational frequencies.

Multi-Frequency Operation

In one embodiment of the present invention, each base station isconfigured to simultaneously monitor at least two frequencies. Eachfrequency may be used to monitor an independent set of transmittersseparated in distance throughout the coverage area for the receiver.Multi-frequency operation of the present invention provides for severalenhancements. For example, an application may be given exclusive use ofa particular frequency for its transmitters. Alternatively, anapplication may be allocated a set of frequencies, where the transmitterselects one of the set for operation. By using multiple frequencies, theoverall data bandwidth of the system can be greatly increased. Asdiscussed above, further advantages are achievable by configuring thesystem to have cells with overlapping coverage areas.

Special problems are caused by the near-far effect due tonon-power-controlled transmitters operating in adjacent channels.Transmitters typically have a spectral rolloff characteristic driven bycomponent and design issues known in the art. As a result, a very strongsignal received in one frequency channel may overlap into an adjacentchannel and impact the usability of that adjacent channel. Typically,adjacent channels are separated by an amount such that the spectralrolloff in adjacent channels is less than the largest anticipateddynamic range. Accordingly, the spectral rolloff will not impact theadjacent channels. Dynamic range is the difference in decibels betweenthe strongest (e.g., nearest) received signal and the weakest (e.g.,farthest) received signal. According to one embodiment of the presentinvention, channel spacing is reduced to levels below that typicallyused for the data rates achievable by the present invention. Channelspacing is minimized by, for example, exploiting the low duty cycle ofthe transmitters used in the system, configuring cells to haveoverlapping coverage areas, and by implementing an abort capability inthe base stations.

FIG. 3 illustrates the near-far problems associated with multi-channeloperation. As shown in FIG. 3, signals HB112 and HB122 are the samesignal received by two different base stations B1 and B2, respectively.Based on the strength of the received signal HB122 at base station B2 ascompared to the strength of the signal HB112 received at base stationB1, it is likely that the transmitter is located closer in proximity tobase station B2 than base station B1. Similarly, signals HB114 and HB124are another signal received from at base stations B1 and B2,respectively.

The top graph HB110 of FIG. 3 shows that the signal HB112 at frequencyf₁ is received at low signal strength at base station B1. Base stationB1 is simultaneously receiving a second signal HB114 at the frequency f₂immediately adjacent in frequency to the frequency Base station B1 maynot be able to successfully receive the signal HB112 at frequency f₁ dueto the overlap in frequency from a strong local transmitter transmittingat frequency f₂.

The bottom graph HB120 of FIG. 3 shows that the signal HB122 atfrequency f₁ is received at high signal strength at base station B2.Base station B2 is simultaneously a second signal HB124 at the frequencyf₂ immediately adjacent in frequency to the frequency f₁. Base stationB2 may not be able to successfully receive the signal HB124 at frequencyf₂ due to the overlap in frequency from a strong local transmittertransmitting at frequency f₁.

Since both base stations B1, B2 are monitoring both frequencies (i.e.,f₁ and f₂) simultaneously, both messages HB122 (HB112), HB114 (HB124)are successfully received by the network. Furthermore, the likelihood ofadjacent channels simultaneously receiving transmissions as illustratedin FIG. 3 is low due to the extremely low duty cycle of thetransmitters. Furthermore, this likelihood is further reduced based onthe fact that typically, only a small number of transmitters aregeographically located such that their transmissions are received asexcessively strong (assuming somewhat uniform distribution oftransmitter placement).

Accordingly, network throughput is maximized by using minimum channelspacing, enabled by low duty cycle transmitters, redundant channels,overlapping coverage areas for base stations, and a data concentrator tocollect the messages from the locationally disperse base stationreceivers.

Auto Frequency Distribution and Channel Loading

In another embodiment of the present invention, total system databandwidth is maximized by using the multi-frequency capacity of thesystem. FIG. 4 illustrates a technique whereby a transmitter maytransmit on any one of N frequencies. As shown in the flow diagram ofFIG. 4, the transmitter first computes a pseudo-random number at stepHK110 to determine which channel should be used. The transmitter thenselects the associated frequency of operation at step HK120. The processthen proceeds to step HK130 where the frequency is allowed to stabilize.Once the carrier frequency stabilizes, the process proceeds to stepHK140 where the packet is transmitted on the selected frequency. If thepseudo-random number generation process is white, each availablefrequency has the same probability of selection. This will result in ahistogram of use that has a flat distribution, as illustrated in graphHK160 of FIG. 4. Graph HK150 of FIG. 4 illustrates a channelized use ofspectrum that would result in the histogram of graph HK160.

Frequency leveling allows for an increased number of transmitters perunit area. This is an important feature of the present invention becausethe cell coverage areas are large. As the radius of a coverage area fora base station increases, the area of use increases as a function of theequation:

area of use=πr ²

where r is the radius of the coverage area.

Multi-Frequency, Non-Channelized Collision Analysis

In one embodiment of the present invention the transmit device selects afrequency of operation between two limits without regard for setchannelization. In this embodiment, collisions must be resolved. FIG. 5illustrates the various conditions that may occur using this approach.As shown in FIG. 5, discrete frequency channels are not observed. HO10of FIG. 5 illustrates a condition where a collision between two receivedsignals will result in the loss of both signals. HO20 of FIG. 5illustrates a condition where the two received signals are separated byfrequency such that there is no collision between the signals, and bothsignals are successfully received. HO30 of FIG. 5 illustrates acondition where one of the received signals is sufficiently strongerthan another received signal. In this case, the stronger signal will besuccessfully received, resulting in the loss of the weaker signal. Thissystem design creates additional channel capacity by utilizing channelspacing that is smaller than the width of the signal bandwidth plus thewidth of the adjacent guardbands. By varying the frequency used fortransmissions, a statistical message loss caused by collisions may beovercome by redundantly transmitting messages. A probability ofsuccessful reception is given by the equation:

Ps=1−[1−e ^(−(2λNT/P)]^(M)

where:

Ps=Probability of a successful reception at a base station;

P=signal bandwidth/available system bandwidth;

=1/time between transmissions;

N=number of remote endpoints within range of the base station;

T=time duration of a data packet; and

M=number of times a single message is redundantly transmitted.

The above discussion is applicable to a single base station receivercollecting data from a dispersed set of endpoint devices. If the basestations are in an overlapped configuration, system throughput can besignificantly improved. A message that was disadvantageously receivedwhile in a collision state and therefore dropped by one receiver will beobserved as a stronger signal by another base station receiver, eachbase station receiver using an abort algorithm as discussed above tocollect both messages from a system perspective.

The introduction of a second or more base station exponentially improvessystem probability of reception for any given transmitter. Theprobability of successful packet reception from a system perspectiveincreases according to:

Ps=1−[1−e ^(−(2λNT/P))]^(M)*^(B)

where:

Ps=Probability of a successful reception of a given transmitter;

P=signal bandwidth/available system bandwidth;

X=1/time between transmissions;

N=number of remote endpoints within range of the base station;

T=time duration of a data packet; and

M=number of times a single message is redundantly transmitted.

B=number of base stations which are monitoring the transmitter

The “B” term above exponentially impacts the ability of the system toreceive the transmitter since it operates as a multiplication functionto the message redundancy index (as shown in FIG. 14).

Available bandwidth may further optimized, for example, by sendingtransmissions at a pseudo-random rate, or sending transmission on pseudorandom-carrier frequencies that are spaced narrower than the signalbandwidth in order to evenly distribute signal occupancy in theavailable spectrum.

Overlapping vs. Non-Overlapping Cellular Coverage Areas

FIG. 6 illustrates a conventional approach to geographically dispersinga set of base station receivers to achieve optimal cellular coverage.Base stations HC10, HC15, HC20, HC25, HC30, HC35 and HC40 are locatedsuch that their areas of coverage do not overlap. Two-way communicationssystems typically provide for control communications where the transmitdevices are power-controlled by the receivers so as to minimize anyimpact on adjacent cells that are not being used.

Techniques for configuring cellular coverage areas are know to those ofordinary skill in the art, and are described in Webb, W., “The CompleteWireless Communications Professional: A Guide for Engineers andManagers,” ISBN 0-89006-338-9, Artech House Publishers, 1999, the entirecontents of which is incorporated herein by reference.

Each particular transmitter HC50 is intended to be observable only to asingle cell base station HC25 which is allocated a specific frequencyf₁. Adjacent cells are allocated other frequencies in order to alleviateinterference from transmitters operating in nearby cells. The pattern offrequency allocation typically requires that several cells separate anytwo cells allocated the same frequency. In some systems dozens offrequencies are utilized to achieve cellular coverage.

Conventional systems also make use of power control to help manage thefrequency reuse issue. A transceiver is commanded by a two-way controlcommunication. Transmit-only applications are configured with a poweroutput setting. In either case, the system must limit the range of thetransmitted signal by controlling the power of the transmission in orderto minimize any impact to adjacent cells.

Additionally, mobile transmitters in a conventional cellular system mustbe managed and handed off from cell base station to cell base station asthe mobile transmitter transitions over cell boundaries.

In one embodiment of the present invention, the system operates in a nonoverlapping cellular structure as shown in FIG. 6. However, many of theproblems and complexities associated with conventional cellular systems(e.g., power control and handoff) are avoided by the system of thepresent invention. Since the duty cycle is low for the transmitters, alltransmitters can use the same frequency. Transmissions entering adjacentcells can be ignored or accepted (see discussion above, regardingoverlapping cells). Further, power control is not necessary since thebase station receivers operate with an extended dynamic range. Also,transmitters within a cell can operate at multiple frequencies, allowingfrequency use to extend system data bandwidth, as compared toconventional systems where multiple frequencies are required in order tolay out the cells geographically in such a way that the system will beoperable.

In one embodiment of the present invention, adjacent cells are designedto ignore transmissions not intended for them by assigning eachtransmitter an integral identification tag that is bound atcommissioning. This approach allows the system to be operated as if itwere a non-overlapping system, even if some overlapping exists.Collision events that impact system throughput can be mitigated usingmessage repetition whereby each message is transmit at least twiceseparated in time by a pseudo random interval.

FIG. 7 illustrates some of the benefits of having overlapping cellsaccording to one embodiment of the present invention. As shown in FIG.7, cell base stations HD 10, HD15, HD20, HD25, HD30, HD35 and HD40 areeach separated in distance at the range of operation of each basestation. As such, each site has an area of coverage that overlaps all ofits adjacent cell base station regions. All base stations may operate onthe same frequency or, alternatively, they may simultaneously operate ona shared set of frequencies. Transmitter HD50 is observable by basestations HD10, HD20 and HD25. Any problems with reception that may beexperience by any one of these base stations are compensated for by theredundant coverage.

In either the overlapped or non-overlapped configurations according tothe present invention, mobile telemetry enjoys a much simpler hand-offor control mechanism. Since the frequency of transmission need not becontrolled, and the power of transmission need not be controlled, thesystem can successfully collect data from the mobile without reverse(command) communication. This embodiment of the present inventionprovides for a mobile transmit-only endpoint that is inexpensive andsimpler to construct and operate than conventional mobile systems.Accordingly, a transmit-receive capability of the mobile device becomesa luxury instead of a necessity.

The present invention extends the state-of-the-art in cellular packetdata communications. Previous cellular packet data communicationsystems, such as those described in Rouquette et al. (U.S. Pat. No.5,920,589), Naden, et al. (U.S. Pat. No. 5,999,561), and Sanderford etal. (U.S. Pat. No. 6,111,911), are also directed to cellular packet datacommunications. However, the present invention provides differentcapabilities and is applicable to a wider range of packet datacommunication applications.

First, the conventional cellular packet data communication systemoperates at a reduced output power relative to the transmitters of thepresent invention, primarily due to regulatory restrictions. Second, thecellular coverage of conventional cellular packet data systems istypically not overlapping, due to the overwhelmingly and prohibitivelyexpensive total count of receivers necessary to cover a metropolitanservice area. Figure element HD60 represents a cell according to aconventional cellular packet telemetry system. A cell according to thepresent invention (e.g., cell covered by base station HD10 in FIG. 7) ismuch larger. In fact, as many as 20 cells must be laid adjacent to eachother in order to achieve a linear coverage equal to a radius of thecell of the present invention. In order to achieve the full coverageprovided by a cell according to the present invention, the total areamust be mapped with the smaller sized cells. In the example used above,this would require up to 400 of the smaller sized cells to achieve thecoverage area of a single cell according to the present invention.Accordingly, it can be appreciated that the total number of cellsrequired to cover a large metropolitan service area enters into thethousands for the conventional cellular packet telemetry systems.Furthermore, overlapping cells of such small coverage area creates avery expensive luxury that is not economically feasible for alarge-scale installation. The sheer task of data concentration from thethousands if not tens of thousands of receivers makes overlapping ofconventional packet telemetry systems untenable. Conversely, the presentinvention can cover even large metropolitan areas with only dozens ofcell towers, representing a much easier data concentration task whichcan be accomplished in a variety of cost effective and availablemethods. Moreover, gaining the benefits of overlapping cells may beachieved at a much lower cost.

According to one embodiment, the present invention uses commerciallyavailable communication towers for a smaller number of cell basestations that cover larger areas. Conventional cellular packet telemetrysystems typically locate their receivers at a lower elevation, such as,for example, on utility poles or other topologies, and often cover onlya subset of a neighborhood or community. The added elevation provided bya communication tower as compared to a utility pole, the antenna gainaffordable due to the significantly smaller numbers of cell basestations required, the increase in output power, and the improvedreceive sensitivity all add up to achieve a far superior range ascompared to conventional cellular packet telemetry systems. By achievingthis extended range, the present invention may be used in manyapplications that are not possible or practical using conventionalcellular packet telemetry technology. The technical benefits of thepresent invention are achieved while remaining an order-of-magnitudeless-costly to deploy and maintain. Furthermore, the desirableperformance characteristics associated with redundant coverage is anaffordable option with system of the present invention.

Error Correction, Sequence Number and History of Data

The system of the present invention implements methods that increase theprobability of successful data throughput. Data reliability andthroughput are improved through a variety of techniques. FIG. 8illustrates a timeline that demonstrates the principle behind thesetechniques. As shown in FIG. 8, each message contains a preamble sectionat the start of the data packet. The preamble portion allows for thereceiver to acquire the signal prior to start of data content. In someembodiments of the present invention, each transmitted message mayinclude a data field that uniquely identifies the endpoint device.

FIG. 9 illustrates the use of data history to improve data integrity inthe system. Conventional cellular packet telemetry systems aresusceptible to data loss by losing a transmitted packet. As shown inFIG. 9, figure elements HY105, HY110 and HY115 correspond to threesuccessive messages which are sent by a remote endpoint device in aconventional cellular packet telemetry system. Each packet (message)includes a packet count and a data payload section (telemetry sample).In the example shown in FIG. 9, packet HY110 is unsuccessfully receiveddue to, for example, interference, collision, or some other reason, andthe data payload is lost forever. Accordingly, the probability ofsuccess for a packet in a conventional system is equal to theprobability of successfully receiving a packet at a receiver.

In one embodiment of the present invention, redundant data is sentwithin each packet to overcome the problem discussed above in regard toconventional systems. Each transmitted message has a sequence numberfield that allows for the receiving system to detect a missed orredundant message. Each data message includes additional payload toinclude unique or redundant data. By transmitting redundant data, ahistory of endpoint activity can be determined, since each messageincludes information related to what has recently transpired, but also,information about what had previously transpired and had already beenreported. For example, one method employed in the present inventionprovides for each message to contain a present reading of a device (suchas a voltage, current or other parametric value) appended by a datafield showing any changes since the last reading or readings insuccession.

FIG. 9 illustrates the principle of redundant data within a message toovercome collision loss according to one embodiment of the presentinvention. As shown in FIG. 9, elements HY120, HY125, HY130 and HY135correspond to four successive messages sent by a particular remoteendpoint device. In the example shown in FIG. 9, message HY125 is lostdue to, for example, interference, collision, obstruction, or some otherreason. Each packet contains a current telemetry sample (present datapayload) plus difference terms for history. The history terms aretransmitted with each message with the number of terms set according tothe requirements for the particular system. In the example, it can beseen that the lost message HY125 can be reconstructed by any subsequentmessage that had a history term corresponding to the packet that waslost, as identified by a packet counter field. Using this technique ofthe present invention, the probability of packet loss is dramaticallylowered. In one embodiment of the present invention, overall messagelength is reduced by compressing the history data using knowncompression techniques.

In another embodiment of the present invention, each data messageincludes an error detection field such as, for example, a cyclicalredundancy check (CRC) field that is used to detect total message biterrors. The system receiver may use this error detection field toprovide error correction as well. FIG. 10 illustrates a method for errorcorrecting a single erroneous bit received in the message according tothe present invention. Message HU110 corresponds to a message with asingle bit error received. The receiver can alternatively toggle asingle bit, rechecking the CRC checksum at each iteration, as shown inHU115 and HU120. Eventually, the correct bit is found, and the CRCchecksum will match, yielding a corrected message HU125, which then isadded to the valid message database.

Using Sequence Numbers to Improve Data Security

In one embodiment of the present invention, embedded sequence numbersare used to enhance system security. This technique is helpful indealing with, a situation where two endpoint devices have the sameunique identifier. This situation could happen by accident or forillicit reasons. By including an auto-incrementing field within eachdata package, this situation can be detected, thereby further improvingdata integrity. FIG. 11 illustrates an example of detecting an illegalendpoint according to one embodiment of the present invention. As shownin FIG. 11, a series of messages HV110, HV115, HV120, HV125, HV130 isreceived. Each message includes an endpoint identification field as wellas an auto-incrementing field. By monitoring the received transmissions,the system detect an illegal transmission as one having a valid (albeitredundant) endpoint identification field with an unexpected value in theauto-incrementing field, as shown in message HV125. The receiving systemcan therefore detect and prevent data from being entered into thedatabase that would otherwise appear valid.

Data Concentrator Function and Operation

In one embodiment of the present invention, the base stations aregeographically dispersed such that a large metropolitan area of hundredsof square miles may be covered with a relatively small number of basestations. In conventional cellular packet data systems, base stationsare positioned roughly ½ mile apart. The system of the present inventionenjoys a 10 mile nominal separation due to frequency of operation andelegance of design. The relatively small number of base stations makesan infrastructure feasible to support a data concentrator that performsthe function of data reduction for data collected by the system. FIG. 12is a flow diagram for a data concentrator process for removing redundantdata collected by the system. The data concentrator is shown as elementHA60 of FIG. 1. As show in FIG. 12, the process begins at step HG10where the data concentrator receives a message from a connected basestation and inserts the message into a database. The process thenproceeds to step HG20 where the data concentrator compares the newlyreceived message with all messages already stored in the database usingan integral identification tag included with each message. The processthen proceeds to step HG30 where any duplicated messages detected usingthe message identification tag are reduced to a single message forstorage. The process repeats to ensure that each received message isonly stored once.

Sectored vs. Omnidirectional Antennas

Conventional cellular radio systems typically become overloaded quickly,primarily due to the predominant mode of continuous operation. FIG. 13illustrates a conventional sectored cell base station. As shown in FIG.13, each cell is sub-divided into three sectors (HI120, HI121, HI122)with associated equipment (HI130, HI131, HI132) to process each sector.Sectoring, therefore, represents an additional cost component that isnot necessary in the present invention.

The present invention deploys a cellular structure for the expresspurpose of collecting packet telemetry of low duty cycle data. Thepresent invention uses an omnidirectional antenna as depicted elementHI210 in FIG. 14. Accordingly, only one antenna and one associatedprocessing receiver HI230 are required to cover an area which typicallyhas twice the radius of conventional cellular radio communicationtechnology.

Base Station Transceiver Capability

So far, only one-way communications (i.e., from the endpoints to thebase station) has been discussed. In another embodiment of the presentinvention, a transceiver is used at the base station allowing the basestation to not only receive packets from the endpoints, but also to sendreverse or outgoing data telemetry from the base station to the remotedevices without impacting the ALOHA channel capacity of the network.

FIG. 15 illustrates a base station with a transceiver according to thepresent invention. The base station has a single antenna HJ110 capableof receiving or sending data. The station antenna HJ110 is connected toan antenna switch HJ115. The antenna switch HJ115 selects between areceiver section HJ120 and transmitter section HJ125 of the transceiver.The base station operates in half-duplex mode to reduce cost and systemcomplexity. The receive/transmit decision is made by a processor HJ130,which communicates with the Data Concentrator HJ160 through acommunications link, such as, for example, a land-line HJ140, a wirelessRF or infrared link, or other communications link. The system ALOHArequirements dictate that out-bound transmissions to remote two-waydevices are restricted to less than 1% duty cycle. This 1% value mustthen be added to the ALOHA system channel capacity which will have aminimal system impact as given by:

Ps=1−[1−e ^(−(λNT+1%))]^(M)

where:

Ps=probability of successful reception at a base station;

=1/time between transmissions;

N=number of remote endpoints within range of the base station;

T=time duration of a data packet; and

M=number of times a single message is redundantly transmitted.

Accordingly, as long as the base station transmitter behaves as a systemendpoint device, having a low duty cycle and transmitting in a packetform, the overall system capacity is not impacted significantly. In oneembodiment, the remote endpoint device targeted for reception uses adata identifier field to filter out unintended messages from alternativetransmitters.

By binding the endpoint devices to a particular base station, the systemon-air traffic may be lowered. A singular cell base station transmittingis preferable to all base stations blindly transmitting to all endpointdevices within a coverage area.

If the remote endpoint device also has transmit capability, the systemcan determine which base station to bind that endpoint device to basedon received signal strength of the endpoint as observed by the systembase station. The base station with the largest received signal strengthmatching the identification tag of the remote transceiver is assignedthe task of transmitting to that endpoint device into the future.

Conventional cellular systems must actively hand-off mobiletransceivers, with the base stations cooperating and allocatingfrequencies and time slots for the hand-off to occur. The system of thepresent invention does not require a traditional hand-off since the basestation receivers are oblivious to the changing location of thetransmitter. In one embodiment, signal strength received at thereceiving stations is used by the data concentrator to select a specificreverse base station at the time of transmission. This serves toautomatically track mobile transceivers through a data network withoutthe need for hand-off by the cell base stations. In another embodiment,a history of throughput is used to select a base station for reversechannel use since each receiver that can receive from the remotetransceiver sends the data to the data concentrator.

Mobile Communication Services in the Cellular Structure

In another embodiment of the present invention, the remote endpointmobile device is equipped with the capability to geographically locateitself, for example, through a global positioning system (GPS)capability. Since the endpoints are aware of their location, systemloading and reliability can be further improved. FIG. 16 illustrates aremote endpoint HA40 with a GPS module HF140 according to the presentinvention. Alternatively, any location device including, but not limitedto an inertial navigation system, a stellar system or any other devicefor determining a geographic position may be used. The transmitter usesthe location information (e.g., latitude, longitude, etc.) to determinewhich frequency to use. This determination is made based on a query of adatabase HF160 using the present location of the transmitter.Transmission is initiated by a sensor HF170 (such as a water meterHS120), an external event HF180 or other wake-up event that is managedby the processor HF150. The sensor or meter HF170 is coupled to theprocessor HF150 by A/D HF175, and the external event HF180 is coupled tothe processor HF150 by buffer HF185.

The wake-up event may be from an internal timer directing thetransmitter to transmit a message, or may be an external event, eitherdigital or analog. In one embodiment of the present invention, thedevice samples the analog input and either transmits the analog value,the rate of change of the analog value, or another message based on athreshold comparison using the sampled value. The processor HF150responds to the wake-up event by creating a packet for transmission. Afrequency reference HF110 is coupled with either the carrier frequencygeneration block HF115 or other mixer HF120 to create a radio frequencysignal for transmission. The processor modulates the RF carrier with thedata, which is then amplified by an amplifier HF125 before beingradiated by the antenna HF130.

The endpoint device according to the embodiment described above in thecontext of FIG. 16 is capable of handling an automatic frequencyswitchover without the need for higher level cellular direction.Accordingly, the device can operate as a mobile transmit-only devicewhile observing a frequency allocation design for a particulargeographic area.

FIG. 17 shows an exemplary layout of a geographical area HP50 in which avertical line HP40 separates two different operating areas that aregoverned by a licensing body or business objectives. A mobiletransmitter might be required to communicate on a first frequency bandin one area, and a second frequency band in the other area. The mobiletransmitters according to the present invention determine their locationusing, for example, a GPS receiver. Conventional mobile tracking systemsuse either manual control or system-directed control to select whichfrequency band is to be used in which locations. The transmitterendpoint device of the present invention stores an on-board databasefrom which a frequency selection is made based on current locationinformation. This determination is made automatically, therefore nomanual control or system-directed switchover is needed.

FIG. 17 includes a flow diagram describing an algorithm for making afrequency determination based on location information. As shown in FIG.17, the process begins at step HP10 where a transmitter device receivescurrent location information from, for example, a GPS receiver. Theprocess then proceeds to step HP20 where the current locationinformation is used to query a database containing location specificfrequency information to determine an appropriate frequency on which totransmit based on the transmitter device's current location. The processthen proceeds to step HP30 where the transmitter device transmits amessage on an appropriate location specific frequency determined in stepHP20.

Applications for mobile transmitters equipped with the capabilitydescribed in the context of this embodiment of the present inventioninclude, but are not limited to, fleet management services where vehiclelocation may be an inherent payload parameter. Furthermore, transmittersentering undefined areas may automatically suspend operation. Oneexemplary benefit of this feature is that unlicensed use may be avoideduntil the transmitter re-enters a location for which it can determine atransmission frequency to use.

FIG. 18 illustrates another exemplary use of this embodiment of thepresent invention. As shown in FIG. 18, a metropolitan service area(MSA) HPQ30 may be allocated two frequencies for operation within thatarea. In the example shown in FIG. 18, the MSA HPQ30 has been allocated220 MHz and 218 MHz, 220 MHz being used throughout the MSA HPQ30, exceptfor within the areas defined by HPQ10 and HPQ20, where 218 MHz must beused. A mobile transmitter according to the present invention receivesits current position from the GPS receiver HPQ40, and determines whichfrequency to transmit on (i.e., 218 MHz or 220 MHz) based on its currentposition. A mobile transmitter that is licensed to operate in all threeareas will automatically alter its output carrier frequency as ittravels throughout the MSA HPQ30.

Smart Repeater Disclosure

In some telemetry installations, there may be a group of transmitendpoints that are located in a challenging RF environment.Transmissions from these endpoints might not be receivable by any basestation receiver. According to one embodiment of the present invention,a repeater is set up so that transmissions by these endpoints will bereceived by at least one base stations. However, this type ofinstallation may significantly increase the ALOHA load of the system,and negatively impact system throughput. A non-discriminating repeaterrepeats every message it receives, even if the repeated messages weretransmitted by strong transmitters and were received by a base stationwithout the aid of the repeater.

Conventional approaches to solving this problem include encoding theidentification numbers of ‘weak’ transmitters into the memory of therepeater. The repeater will then compare the transmitter ID number ofevery received message against the list of ID numbers stored internal tothe repeater. The repeater will then transmit only those messages thatit received from the approved list of ‘weak’ transmitters. This methodhas a disadvantage that all transmitters must be known when the repeateris installed.

The repeater according to the present invention, however, does notrequire prior information of the ID number of each transmitter whosemessages require repeating. The repeater according to the presentinvention has a special ‘program’ mode that aids in selecting whichtransmitters should have their messages repeated. FIG. 19 illustrates anapproach for dealing with messages that are not received by any basestation according to the present invention. In the example shown in FIG.19, messages transmitted by a ‘weak’ endpoint transmitter HR10 are notbeing received by any base station. In response to this condition, arepeater HR20 is located in close proximity to the endpoint transmitterHR10 and placed into ‘program’ mode. In ‘program’ mode, repeater HR20goes into a receive-only mode for a certain time period, and stores theID numbers (one of which is the ‘weak’ endpoint transmitter HR10) ofeach transmitter from which a message was received into a database. Inone embodiment of the present invention, ‘weak’ endpoint transmitterHR10 is forced to transmit during the time that repeater HR20 is in‘program’ mode. As a result of the programming step, the repeater HR20has stored the ID of the ‘weak’ endpoint transmitter HR10 into itsdatabase. The repeater will then be moved to a permanent installationlocation HR30, which is within range of both the ‘weak’ endpointtransmitter and a base station HR40.

FIG. 19 includes a flow diagram describing the process of programmingthe repeater HR20 according to one embodiment of the present invention.As shown in FIG. 19, the process begins at step HR50 where the repeaterHR20 is placed into programming mode. The process then proceeds to stepHR60 where the repeater HR20 receives messages and measures the signalstrength of each received message. The process then proceeds to stepHR70 where the signal strength of each received message from eachtransmitter ID is compared to an entry from a temporary data base withthe highest stored signal strength from that transmitter ID. The processthen proceeds to step HR80 where if the new signal strength for a giventransmitter ID is stronger than the stored signal strength, thetemporary database is updated with the new value. As shown in step HR90,the process continues for a predetermined interval of time “T.” Thetransmitter ID corresponding to the highest signal strength value (whichshould correspond to the ‘weak’ endpoint transmitter) will then bestored in a permanent database HR110. The above described process isperformed by a processor or other algorithmic device HR100 capable ofperforming the prescribed function.

In another embodiment, the smart repeater HR20 uses a time acceptancewindow exclusively, ignoring the signal strength of the receivedmessages. This approach removes the need to closely locate the repeaterin the program mode. In this embodiment, the time acceptance window isshort, and the ‘weak’ endpoint device is made to transmit within thetime acceptance window. In yet another embodiment, the smart repeaterHR20 is configured to allow for a transmitter ID to be communicated tothe smart repeater HR20, through, for example, a special message formatrecognizable by the smart repeater HR20.

The smart repeater of the present invention is applicable in a varietyof systems, including, but not limited to, narrowband, wideband, directsequence spread spectrum, frequency hop and any other digitalcommunication technology where the transmitting endpoint is identifiableby some sort of identification tag. The identification tag allows therepeater to identify and distinguish messages based on their origin.

Boost Mode

In one embodiment of the present invention, sensitivity that is lost dueto sub-optimal conditions is recovered at both the transmitter and thereceiver. Transmit-only remote endpoint devices are sometimes used inapplications that hinder the device's ability to be received by theintended receiver. In this embodiment of the present invention, the lostsensitivity is recovered without incurring the cost of a higher outputpower of the transmit-only device. This is accomplished on a systemlevel, both at the transmitter and at the receiver.

In this embodiment of the present invention, the system is placed into amode designated as “boost” mode. “Boost” mode dedicates at least one ofthe system channels as a “boost mode” channel. In this mode, the systemoperates as a multi-channel receiver architecture at the base stationwhere each of the channels are independently configurable for use. Thereceiver dedicates at least one system channel for reception of theboost transmitters.

FIG. 20 illustrates typical frequency allocations for both normal andboost channels from the perspective of a multi-channel receiver system.FIG. 20 illustrates a normal, non-boost frequency allocation HH120,HH130. The signal occupying the channel identified as HH120 is shown asbeing strong in signal-to-noise ratio (SNR), and therefore, should bereceived properly. The normal signal occupying the channel identified asHH130, however, is low in SNR and therefore, is a candidate forswitchover to boost operation.

The boost transmitters transmit in a specified boost channel, identifiedin FIG. 20 as HH140, at a significantly lower data rate than the other,normal system channels. The receiver detects boost transmitters in asmaller bandwidth (a sub-boost bandwidth) within the system boostchannel. The noise floor for reception has a mean power proportional tothe receiver detector bandwidth. The gain in sensitivity is10*log(bandwidth reduction ratio). The receiver sub-boost channelbandwidth is about equal to the bandwidth of the boost signal whichproduces a reduced noise floor HH160 resulting in greater link budget.By lowering the baud rate (i.e., data rate), the system provides forbetter sensitivity as shown in the following formula:

SNR improvement=10*log(normal baud rate/lower(boost) baud rate)

=10*log((16.64 kb/s)/(520 b/s))=

15 dB

Some installations attenuate the transmitter radiation by 10-15 dB. Thisresults in a loss of SNR at the receiver and limits the ability to usethe preferred higher data rate techniques of the transmitter. The 15 dBSNR improvement provided by boost mode, as noted above, can overcome theestimated 10-15 dB loss due to this restriction. The present inventionuses this receiver SNR gain to enable transmitter installations inpreviously unusable locations such as buried water meters.

Boost mode not only increases sensitivity at the receive end, but alsocan significantly increase the service coverage area while maintainingexpected battery life and favorable collision statistics, even thoughthe message duration is increased due to the decrease baud rate,providing that the number of transmissions be reduced by the followingratio:

Normal baud rate/lower (boost) baud rate

The service area range extension due to the SNR increase provided for byboost mode, in one example, allows for a 235 square mile extension asillustrated in FIG. 21. In this example, the receiver is placed at acentral location HH210, and the radius of the standard coverage areaR_(N) is equal to 5 miles HH220. Using these numbers, the standard areacovered is:

Normal baud covered area=π*(5)²=79 square miles.

If 15 dB of boost mode SNR improvement extends the usable range to a 10mile radius R_(B) HH230, then the additional coverage provided by boostis:

$\begin{matrix}{{{Additional}\mspace{14mu} {Boost}\mspace{14mu} {coverage}} = {{\pi*(10)^{2}} - {\pi*(5)^{2}}}} \\{= {314 - 79}} \\{= {235\mspace{14mu} {square}\mspace{14mu} {{miles}.}}}\end{matrix}$

In FIG. 21, the extended boost mode coverage area is shown as HH235.Furthermore, since the baud rate has been significantly lowered, thecorresponding bandwidth of the signal has been significantly lowered,thereby allowing for several boost sub-channels to fit within thededicated system channel.

FIG. 22 illustrates a proportionality relationship for the reduction ofpacket transfers in a boost enabled system according to one embodimentof the present invention. As the boost messages are longer in duration,the overall number of packets to be received is reduced in proportion tothe ratio of data rates between boost and normal operation.

An additional benefit of boost mode according to the present inventionis illustrated in FIG. 23. As shown in FIG. 23, the allocated boostchannel is located between two normal system channels. Strongtransmitters channels adjacent to the boost channel will overlap theboost channel. This is caused by, for example, frequency drift,oscillator phase noise, PLL spurs, modulation roll-off, transmitter datafilter roll-off, crystal aging, Doppler shift, or other causes inherentin the transmitter or receiver system components. The impact of theadjacent channel bleed-over can be mitigated by concentrating the boostsub-channels HNN30, HNN32, HNN34 toward the center of the dedicatedboost band. By doing this, the bleed-over of the strong signals HNN21,HNN22 into the boost channel boundaries HNN12 and HNN14 does not impactthe performance of the boost communications.

Due to a fixed frequency error budget for the system according to thepresent invention, the reduced baud rates do not proportionally increasethe number of usable channels, as illustrated in FIGS. 24 and 25. Thefrequency error budget is equal to the fractional system frequency errorbudget times the carrier center frequency. As the carrier centerfrequency gets higher, the required guard band bandwidth can increasebeyond the received signal bandwidth. As shown in FIGS. 24 and 25, innormal, non-boost operation, signal HM30 requires additional guard bandHM40 on either side of the desired center frequency of operation toallow for frequency reference error. Similarly, boost channels HN30 mustalso allow for guard band HN40. However, since guard band is not afunction of signal bandwidth, the proportional band usage ratio is not aconstant. Regardless of the mode (i.e., normal or boost), the guard bandHN40, HM40 required is the same, provided the center frequencies ofoperation are identical.

Reduced baud rates do not proportionally increase the number of usablechannels. This results in a 5× waste of available bandwidth.Furthermore, as the center frequency of operation goes higher, the guardband alone increases, thereby further reducing the ratio of signal toguard bandwidth.

The boost mode SNR gain provided by the present invention can be used torecover data from installed endpoints that was previously unfeasible.FIG. 26 illustrates a few exemplary installation problems that aremitigated by boost mode. The present invention uses boost mode forremote endpoint devices HA40 that are disadvantageously located, forexample, on the side of a building HS 100, or in various buried meteringapplications HS 110. Meter HS 120 represents an underground meteringapplication for the remote endpoint HA40 such as, for example, water,fuel, or flow. The endpoint transmitter is co-located at the mostadvantageous spot possible. In the example shown in FIG. 26, thetransmit device HS 130 is mounted to a meter access port. In thisexample, an antenna HS 140 is integrally built into the meter accessbarrier.

FIG. 27 illustrates a channel utilization map for one embodiment of thepresent invention. In this embodiment, the system is configured forarea-wide packet data. As shown in FIG. 27, boost channels are allocatedto Channel 3 in this example. Channel 10 is indicated as being reserved.In one exemplary embodiment Channel 10 is used by a distant base stationto repeat locally received messages for reception by another basestation. This exemplary use is illustrated in FIG. 1 where a distantbase station HA70 repeats messages that it has received on Channel 10(i.e., f₁₀) to base station B₃ HA30.

Alternatively, the present invention allows a looser definition of theallowable boost channels that may be used, which will allow the centerfrequencies to drift significantly. This mitigates the issue of guardband waste as long as the receiving system has the ability to process aboost channel over a band including the entire band of operationallocated for boost plus the error guard band. Using this approach, thesystem functions more like a randomized transmit frequency system than achannelized transmit frequency system. This is due to allowing forfrequency error of the transmitter which will cause the signal to showup anywhere in the system channel versus in a predicted sub-channel. Inthis mode of operation, the corresponding receiver uses windowed-FFT'sto evaluate the entire system bandwidth as shown in FIG. 50, adescription of which is included herein.

In one embodiment of the present invention, boost sub-channels arenominally separated by a predetermined frequency spacing. In anotherembodiment, a further improvement in the system achieved by ensuring aneven distribution of the usage of the boost sub-channels by having theboost transmitter transmit on all of the sub-channels eithersequentially or in a pseudo-random fashion. In yet another embodiment,the transmit frequency is dithered within the sub-channel spacing. In analternative to dithering transmission frequencies for each transmission,each transmitter could be tuned during manufacturing to one of adistribution of different frequencies within the sub-channel.

Battery Life of Transmit-Only Endpoint Device

As described above, the present invention provides for low cost, batteryoperated endpoint devices that have multi-year operation. A single,readily available battery will provide sufficient energy due to the lowduty cycle of the transmitter, assuming that the transmit endpointdevice has the capability for entering a reduced power state for themajority of its service life.

FIG. 28 illustrates the multi-year operation using a single 1.4 amp-hour(AH) battery. If a 15 millisecond data burst with a 15 minute repetitionrate is assumed, the resulting duty cycle is:

duty cycle=0.015/(15*60)=0.000016666

In one embodiment of the present invention, the transmitter radiates 2Watts (W) of power and typically requires 550 milli-Amperes (mA) foroperation. An average transmit current can be calculated based on theduty cycle and average current as:

Average transmit current=550 mA*0.000016666=9.2 μA/sec

Typically, the transmitter makes a transition from a sleep state to aready-to-transmit state which may take a few milliseconds (e.g., 10 ms)at some lower current consumption, (e.g., 25 mA in one embodiment of thepresent invention). The transmit set-up current is additive andcalculated as:

Transmit set-up current=(0.010*25 mA)/(15*60)=0.3 μA

In one embodiment of the present invention, the current consumption ofthe transmitter in a sleep state is 5 μA in a sleep state:

Sleep current & leakage=5.0 μA

Therefore, a the total average operation current for this examplebecomes:

Total transmitter current/second=9.2+0.3+5.0=14.5 μA

Using a conventional 1.4 AH lithium battery (derated 20% to compensatefor self leakage), the transmitter service life in this embodiment ofthe present invention can be calculated as:

(1.4 AH*0.8)/14.5 μA=72,241 hrs=8.8 years

Accordingly, with update rates in the order of 15 minutes, devicesaccording to the present invention which transmit in the 15 msecduration range can operate on a single battery for multiple years, evenwhile transmitting 2 Watts.

Targetable Software Update

Embedded computer systems are becoming more common in the radio arts,and more than one embedded processor is often used in these systems.Periodically, the software for these systems must be modified toaccommodate new system requirements. The diverse locations of many ofthese base stations dictate that the embedded software be updatedremotely. Standard practice would update software for all embeddedprocessors within a system. Partial software updates, or an update foronly one of several processors within a system, introduces undue systemcomplexity. The present invention provides an approach for performing apartial software update without introducing additional systemcomplexity.

In one embodiment of the present invention, specific portions ofembedded software can be updated remotely. FIG. 29 illustrates a DataConcentrator RJ10 connected to “n” remotely located base stations RJ20,RJ30, RJ40. Each base station RJ20, RJ30, RJ40 is connected to anantenna RJ22, RJ32, RJ42, respectively. Block RJ50 is a block diagram ofa base station. Each base station RJ20, RJ30, RJ40 is connected to theData Concentrator RJ10 by a network connection. Base station RJ50includes at least one central processor RJ55 that is directly accessibleto the Data Concentrator. Processor RJ55 has a dedicated centralprocessor program memory RJ57. Processor RJ55 interfaces with a centralcommunication bus RJ60, whereby processor RJ55 communicates with otherembedded processors in the system. In one embodiment of the presentinvention, the base station receiver uses a plurality of channel modulesto simultaneously search multiple frequency channels. Channel Module 1RJ70 contains a dedicated control channel processor RJ72 connected to achannel program memory device RJ73. Control channel processor RJ72 usescommunication bus RJ74 to communicate with channel subprocessors RJ75through RJ77, each having dedicated memory hardware RJ78 through RJ80.It is not necessary that processors RJ55, RJ72, and processors RJ75through RJ77 are of the same type. Central processor RJ55 usescommunication bus RJ60 to communicate with multiple channel modulesRJ70, RJ80. In one embodiment of the present invention, each of theprocessors in the base station RJ50 has a unique identification number.

In one embodiment of the present invention, a message protocol allowsfor the remote update of a single memory sub-system in the base stationarchitecture described above. Using this protocol, the Data Concentratormodifies individual memory locations of any of the memory deviceslocated at any of the base stations in the system.

FIG. 30 illustrates a message protocol for modifying a memory of a basestation according to one embodiment of the present invention. As shownin FIG. 30, the message RJ210 includes a message header field RJ220, adestination processor identification field RJ230, a memory addresslocation field RJ240 indicating which memory address to update, aprogram memory data field RJ250 containing the program memory data thatwill be stored at the location indicated by the memory address locationfield RJ240 and an error detection data field RJ260 containinginformation used in performing error detection.

LAN-to-WAN Bridge

FIG. 31 illustrates a LAN system being connected to a WAN systemaccording to one embodiment of the present invention. The systemincludes a transmit-only (one-way) LAN subsystem RK110 that has atransmitter RK112 and receiver RK116. The system also includes atransmit-only (one-way) WAN subsystem RK120 that has a transmitter RK122and receiver RK126. The LAN receiver RK116 is coupled to the WANtransmitter RK122 by a bi-directional LAN-to-WAN bridge RK130. The WANtransmitter RK122 is programmed to transmit a message immediately uponreceiving the message via the LAN-to-WAN bridge RK130 from receiverRK116. Alternatively, transmitter RK122 can be programmed to transmitthe received message after a set or pseudo-random delay. In this way, anALOHA-based transmission on one network will generate an ALOHA-basedtransmission on a separate network.

FIG. 32 illustrates an advantageous use of the LAN-to-WAN bridge of thepresent invention. As shown in FIG. 32, an endpoint transmitter RK212 islocated outside of a licensed coverage area RK230 of a WAN. The LANtransmitter RK212 wirelessly transmits messages to the WAN receiverRK216, which in turn transmits the message via the LAN-to-WAN bridgeRK130 to WAN transmitter RK222. WAN transmitter RK222 located within thelicensed coverage area RK230, and will transmit the message on thelicensed frequency band to WAN receiver RK226.

FIG. 33 illustrates another example, where an endpoint transmitter RK312is located in a disadvantageous location with respect to the system WANreceiver RK326. In this case, a repeater may be used to bridge the areaof obstruction to a more advantageous area of reception relative to thesystem WAN receiver RK326. Bridge RK316 transfers messages receivedinside the area of obstruction to a transmitter outside the area ofobstruction, which then retransmits those messages to a receiver RK326.

FIG. 34 illustrates a third exemplary use of the LAN-to-WAN bridge wherethe bridge serves as a LAN concentrator, receiving messages from aplurality of remote endpoint devices RK412, RK414. The bridge RK416 thenrelays all or a selected group of messages from the remote endpointdevices to the system receiver RK426.

Remote Serial Data Monitoring

In one embodiment of the present invention, a system remotely monitorsserial devices as well as sensor components. Remote endpoint devicesserve as a serial to RF translation devices, transforming the serialdata to packet telemetry messages to be broadcast over the system andreconstituted at some other destination into a serial stream.

FIG. 35 illustrates a conventional approach for monitoring serial dataas well as system for monitoring serial data according to one embodimentof the present invention. As shown in FIG. 35, the conventionalcommunication system interfaces a serial device HX10 to a monitoringdevice HX20 through a hard-wire connection. For distributed systemswhere a plurality of remote endpoints are concentrated, thisconventional approach requires that a separate hard-line wire be run toeach endpoint HX30, HX40, HX50 from the monitoring device HX60.Sometimes, the endpoint devices HX30, HX40, HX50 are serially strung tolimit the wire to one physical component.

In one embodiment of the present invention serial devices are eachequipped with an endpoint transmitter, alleviating the need to hardwireall of the monitored devices together. Moreover, the monitoring deviceneed not be co-located with the devices being monitored, and cantherefore be placed at a convenient location. As shown in FIG. 35,serial devices HX70, HX80 are provided radio transmitters HX90, HX100which serve as the remote endpoint devices in the wireless network. Abase station or group of base stations HX110, HX120 receive messagesfrom the remote transmitters. A data concentrator HX130 performs datareduction and presents the data to a device monitoring data concentratorHX140 that may be placed in any convenient location, even miles orcontinents away from the devices being monitored. In one embodiment ofthe present invention, the link between the data concentrator HX130 andthe device monitoring data concentrator HX140 is the Internet. In anembodiment where the link to device monitoring data concentrator is theInternet, the device monitoring data concentrator HX140 may in fact bemany devices, widely separated and uncoordinated. The serial signal fromthe serial telemetry devices HX70, HX80 is reconstituted at a serialoutput telemetry device monitoring device HX150 or at as many endpointsas is desired.

Transmitter Device

FIG. 36 illustrates a typical FM RF architecture that accepts amodulating waveform, generates a modulated carrier, amplifies it, andpresents it to an antenna. FM modulation and RF amplification techniquesare well known in the digital communication art, and are therefore, notdescribed in detail herein. Using FM allows for more power efficient(i.e., non-linear) power amplifiers than non-constant envelopemodulation methods.

As shown in FIG. 36, the architecture includes a frequency source E100,a frequency modulator E150, a power amplifier E170, an antenna E180, anda modulating waveform generator E105. The frequency source E100 is acrystal-controlled frequency synthesizer, or any other frequency source.The modulating waveform generator E105 includes a leader patterngeneration block E110, a message source E120, an appending block E130,and a modulating symbol waveform generator E140.

The leader pattern generation block E110 generates a leader, which is arepeating pattern of bits that are appended to each data packettransmitted, and are used by the receiver for signal detection andacquisition processing. The message source E120 provides the informationbits. The appending block E130 appends each data message from themessage source 120 with a leader from the leader pattern generationblock E110 to create the complete sequence of bits to be transmitted.The output of the appending block E130 is provided to a modulatingsymbol waveform generator E140.

The frequency modulator E150 receives the modulating symbol waveformfrom block E140 and the carrier waveform generated by the frequencysource E100 as input. The frequency modulator E150 modulates the carrierwaveform with the modulating symbol waveform to produce a frequencymodulated carrier wave that is amplified by the power amplifier E170 andcoupled to the antenna E180.

In a more complex FM transmitter architecture, a power management unitE190 is also included. The power management unit E190 includes anultra-low power timekeeping unit. The timekeeping unit provides power tothe other elements of the transmitter at predetermined time(s), therebyconserving power when transmissions are not required. This powerconservation technique requires that the frequency source E100 be ableto start rapidly when power is applied. Rapid starting crystaloscillators are readily available devices.

In one embodiment of the present invention, the frequency source E100and frequency modulator E150 include a crystal oscillator and a phaselocked loop (PLL) that includes a high pass frequency modulation input.This is a common design that has a relatively low cost. Typically, inorder to pass the modulating symbol waveforms with full fidelity, thePLL time constants are 5, 10, 20, 50, 100, or more times longer than afrequency shift keying (FSK) symbol duration. In a transmitter wherepower management was important, these PLL time constants wouldsignificantly degrade battery life due to the long start-up or frequencychange settling time required by the PLL.

A modulation transfer function (MTF) is a broad bandpass transferfunction that requires that the modulating waveform contain no DCresponse, and no low frequency spectral component. In conventional M-aryFSK, the information bit stream is encoded to remove the DC response andthe low frequency spectral components. An impact of this encoding is areduction in the effective bit rate.

In one embodiment of the present invention, the PLL time constants areset to be about equal to the FSK symbol rate. Shorter PLL time constantsconserve energy and extend battery life. However, having short timeconstants for the PLL results in a bandpass MTF that rolls off steeply(−40 dB per decade) below the symbol rate. This degradation is correctedat the receiver with an equalizing filter B282 in FIG. 54 that followsthe frequency demodulator. Since the noise spectrum at the output of thefrequency demodulator in the receiver rolls off steeply with decreasingfrequency, the equalizing filter B282 in FIG. 54 causes no significantdegradation of receiver sensitivity.

In one embodiment of the present invention, the leader pattern generatedby the leader pattern generation block E110 is kept relatively short.Having a short leader pattern will help to conserve transmitter powerand extend battery life of the transmitter. A rapid signal detection andacquisition process used by the receiver enables the use of a shortleader pattern.

The transmitter leader must be long enough for the receiver tosuccessfully acquire the message. If the receiver has a priori knowledgeof the transmitter frequency, the acquisition will only require theleader length to be as long as the worst case delay in the detectionprocess. In one embodiment of the present invention, the FM receiver mayoperate with limited sensitivity and require a leader being only one ortwo symbols in length. The receiver of the present invention implementsaveragers and filters with group delay, which require the leader have alength of between 16 and 32 symbols to achieve optimal sensitivity. Inyet another embodiment, where the transmitters are frequency agile andthe receiver must scan multiple frequencies to search for thetransmission, the leader must be lengthened to provide time for thereceiver to acquire the signal. Depending on the number of frequenciesbeing monitored, the leader can grow to up to 60 to 100 symbols inlength.

FIG. 37 illustrates a novel approach to encoding a message bit streamaccording to one embodiment of the present invention. This approachremoves the DC response and low frequency spectral components whilepreserving the bit rate. The encoder shown in FIG. 37 corresponds to themodulating symbol waveform generator block E140 in FIG. 36. As shown inFIG. 37, the encoder is a 16QAM submodulator. Four bits are encoded intoeach symbol as shown in the QAM constellation E200. Accordingly, theresulting symbol rate is ¼ the bit rate. The QAM submodulator uses aconventional QAM modulator with Walsh carrier basis functions E220,E230, and has a carrier frequency equal to the symbol rate. QAMmodulators are known to those of ordinary skill in the digitalcommunications art and are described in Rappaport, T. S., “WirelessCommunications: Principles and Practice,” ISBN 0-13-375536-3, PrenticeHall, 1996, the entire contents of which is incorporated herein byreference.

According to the present invention, the output of the 16QAM submodulator(i.e., encoder) is a 7FSK modulating waveform E290. The 7FSK modulatingwaveform is made up of a sequence of 4 bit symbols. Each of the 4 bitsof a symbol corresponds to 1 of 7 frequencies of the frequency modulatorE280. The frequency modulator E280 will send the frequency correspondingto each bit of the symbol for a 1 bit period. 7FSK has the samebandwidth efficiency E210 and power efficiency as 4FSK if the peakdeviation of both 7FSK and 4FSK is the same. The 7FSK symbol waveformscan be implemented as entries of a look-up table addressable by the 4bits defining a particular symbol. Accordingly, the 16QAM encoder may beimplemented with very low cost logic (e.g., a general purposemicrocomputer).

In one embodiment of the present invention, the corresponding receiveruses an incoherent frequency demodulator. The use of an incoherentfrequency demodulator causes a 3 dB sensitivity loss to be incurred ascompared to using a coherent demodulator. In one embodiment of thepresent invention, the peak deviation of the 7FSK is increased by afactor of 1.5 times the peak deviation of a coherent 4FSK. This increasein peak deviation increases the sensitivity of the correspondingreceiver by 3.5 dB. Expanding the peak deviation will also increase theRF signal bandwidth by a factor of 32/25, which is an acceptabletradeoff for the transmitter of the present invention. Accordingly, theincoherent receiver of the present invention can achieve a bit errorrate (BER) approximating a BER of a coherent system.

PWM Synthesizer

FIG. 38 illustrates a fractional-N phase locked loop (PLL) frequencysynthesizer. As shown in FIG. 38, the frequency synthesizer includes aninput reference frequency F_(ref) E300, a phase frequency detector E310(PFD), two loop filters E320, E340, a voltage controlled oscillator(VCO) E360, a dual-modulus divider E380 and digital logic E390 tocontrol the dual-modulus divider E380. The synthesizer produces anoutput frequency F_(out) E370. Conventional frequency synthesizersimplement the digital logic E390 with counter logic devices.Fractional-N phase locked loop frequency synthesizers are know to thoseof ordinary skill in the digital communications art, and are describedin Rohde, U. et al., “Communications Receivers: Principles and Design,”Second Edition, ISBN 0-07-053608-2, McGraw-Hill, 1997, the entirecontents of which is incorporated herein by reference.

In one embodiment of the present invention, the digital logic E390 isimplemented as a pulse width modulator (PWM) to control the dual-modulusdivider E380. The output frequency of this synthesizer isF_(out)=F_(ref)(N+FREQ/L) where N is the divider modulus, FREQ is abinary number setting the output frequency and L is the number of statesin the PWM. Pulse width modulators are well known in the art, therefore,detail is not provided herein. Pulse width modulators are readilyavailable, and inexpensive components.

In one embodiment of the transmitter, N=64, L=68, and FREQ ranges from 0to 68 decimal and is retrieved from a table of stored operatingfrequencies E395. In this embodiment, the PWM is a component of ageneral purpose microcomputer. To obtain adequate spurious responselevels from the synthesizer, notch filters E330, E350 are added to thePLL loop filters E320, E340. Filters E320, E330, E340 and E350 may bearranged in any order to accomplish the same result. In this embodiment,the notch frequency of filter E330 is twice the PWM frequency, and thenotch frequency of filter E350 is equal to PWM frequency. In order tominimize cost, notch filters E330 and E350 are implemented as RC twin-Tfilters. To maintain PLL stability, notch filter E350 is an activefilter having a higher Q than the passive RC twin-T filter E330. The PLLfiltering techniques and notch filter design techniques described aboveare well known in the art; therefore, detail is not provided herein.

Theta-r Trip and Signal Acquisition Algorithm

FIG. 39 illustrates one embodiment of a signal acquisition algorithm ofa receiver for acquiring a signal transmitted using the M-ary FSKmodulation with QAM submodulation described above. As shown in FIG. 39,a FM demodulator block A110 translates a received time domain signalinto a modulation domain signal. A QAM Trip Correlator A115 receives themodulation domain signal and generates I and Q data signals that areinput to an averager A120 to produce averaged I and Q data signals. Theaveraged I and Q data signals are input to a Cartesian-to-Polarconverter A125 to generate r (magnitude) and Θ (angle) values. Themagnitude term is proportional to the frequency deviation, the angleterm is the relative symbol phase of the transmitter with respect to thereceiver.

At each symbol time n (i.e., n=0, 1, 2, . . . ), the magnitude valuer(n) is passed to a delay element A130, to produce a ‘previous’magnitude value r(n−1) for the next symbol. A proximity detector (i.e.,a comparator) A135 compares the absolute value of the difference betweensignals r(n) and r(n−1) to a predetermined threshold value. If theabsolute value of the difference between r(n) and r(n−1) is less thanthe predetermined threshold value, the comparator generates a TRUE(i.e., ‘1’) value at the output of A135. If the absolute value of thedifference between r(n) and r(n−1) is greater than the predeterminedthreshold value, the comparator generates a FALSE i.e., ‘0’) value atthe output of A135.

At each symbol time n (i.e., n=0, 1, 2, . . . ), the 0 value Θ(n) ispassed to a delay element A140, to produce a ‘previous’ Θ value Θ(n−1)for the next symbol. A proximity detector (i.e., a comparator) A145compares the absolute value of the difference between signals Θ(n) andΘ(n−1) to a predetermined threshold value. If the absolute value of thedifference between 0(n) and Θ(n−1) is less than the predeterminedthreshold value, the comparator generates a TRUE (i.e., ‘1’) value forthe output of A145. If the absolute value of the difference between 0(n)and Θ(n−1) is greater than the predetermined threshold value, thecomparator generates a FALSE (i.e., ‘0’) value for the output of A145.

The outputs of the two proximity detectors A135, A145 are provided to anAND block A150. The trip algorithm is satisfied if both inputs to theAND block A150 are TRUE. If a value of the output of either proximitydetector A135, A145 is FALSE, the trip condition is not met, and thereceiver will continue its trip algorithm processing, searching for avalid received signal.

The trip algorithm according to this embodiment of the present inventionmakes use of a transmitter leader. In this approach, a known, fixedsymbol pattern is transmitted for at least the acquisition time of thereceiver. Since the symbol rate of the receiver matches the symbol rateof the desired transmitter, and the transmitter repeats the same symbolpattern as the leader of each transmission, the receiver can expect toobserve the same relative angle and magnitude over subsequent symbols ofthe leader, and thereby acquire the signal.

FIGS. 40 through 46 illustrate another embodiments of a trip algorithmaccording to the present invention. In FIG. 40, blocks A210, A215, A220,and A225 duplicate the functions performed by blocks A110, A115, A120,and A125 shown in FIG. 39, respectively.

As shown in FIG. 40, additional delay elements are added such that ateach symbol time n (i.e., n=0, 1, 2, . . . ), the r value r(n) is passedto a first delay element A230, to produce a ‘previous’ r value r(n−1)for the next symbol, the r(n−1) is then passed to a second delay elementA231 to produce a ‘twice delayed’ r value r(n−2). Similarly, the 0 valuetheta(n) is passed to a first delay element A240, to produce a‘previous’ 0 value Θ(n−1) for the next symbol, the 0(n−1) is then passedto a second delay element A231 to produce a ‘twice delayed’ 0 valueΘ(n−2).

FIG. 43 illustrates the generation of a ‘narrow magnitude trip’ signalaccording to one embodiment of the present invention. As shown in FIG.43, the three input signals r(n), r(n−1), and r(n−2) correspond to thesignals created by the delay elements in FIG. 40, discussed above. Thesethree signals are input into averaging block A310 to produce an averagemagnitude value. A ratiotic acceptance constant A330 is multiplied by amultiplier A320 with the average magnitude value from the averagingblock A310 to produce a limit differential. The limit differential issummed with the average magnitude value in a summer A340 to produce aNarrow Upper Limit (NUL) signal. A difference between the limitdifferential and the average magnitude value is determined by adifference block A350 to produce a Narrow Lower Limit (NLL) signal. Acomparator block A360 compares the input signals r(n), r(n−1), andr(n−2) with the NUL and NLL signals. If the values of the three inputsignals r(n), r(n−1), and r(n−2) are all between the NUL and NLL values,then the value of the ‘narrow magnitude trip’ signal is TRUE (i.e.,‘1’). If any of the signals r(n), r(n−1), and r(n−2) is not between NULand NLL, then the value of the ‘narrow magnitude trip’ signal is FALSE(i.e., ‘0’).

FIG. 44 illustrates the generation of a ‘wide magnitude trip’ signalaccording to one embodiment of the present invention. As shown in FIG.44, the three input signals r(n), r(n−1), and r(n−2) correspond to thesignals created by the delay elements in FIG. 40, discussed above. AWide Upper Limit (WUL) value is stored in block A410, and a Wide LowerLimit (WLL) value is stored in block A430. A comparator block A420compares the three input signals r(n), r(n−1), and r(n−2) with the WULand the WLL values. If the values of the three signals r(n), r(n−1), andr(n−2) are all between the WUL and WLL values, the value of the ‘widemagnitude trip’ signal is TRUE (i.e., ‘1’). If any of the signals r(n),r(n−1), and r(n−2) are not between WUL and WLL, the value of the ‘widemagnitude trip’ signal will be FALSE (i.e., ‘0’).

FIG. 45 illustrates the generation of a ‘theta/angle trip’ signalaccording to one embodiment of the present invention. As shown in FIG.45, the three input signals Θ(n), Θ(n−1), and Θ(n−2) correspond to thesignals created by the delay elements in FIG. 40, discussed above. A Θthreshold value is stored in block A510. Block A520 determines anabsolute value of the difference between (1) Θ(n−1) and Θ(n); (2)Θ(n−2), and Θ(n); and (3) Θ(n−2), and Θ(n−1). Block A520 compares eachof these differences with the Θ threshold. If the absolute values of thedifferences calculated by Block A520 are all less than the Θ thresholdvalue, the value of the ‘theta/angle trip’ signal is TRUE (i.e., ‘1’).If any of the absolute values of the differences calculated by BlockA520 are not less than the Θ threshold value, the value of the‘theta/angle trip’ signal will be FALSE (i.e., ‘0’).

FIG. 46 illustrates the generation of a combined signal acquisition(trip) signal according to one embodiment of the present invention. Asshown in FIG. 46, the ‘narrow magnitude trip’ input signal correspondsto a narrow magnitude trip signal described above in the context of FIG.43, the ‘wide magnitude trip’ input signal corresponds to a widemagnitude trip signal described above in the context of FIG. 44, and the‘theta/angle trip’ input signal correspond to a theta/angle trip signaldescribed above in the context of FIG. 45. As shown in FIG. 46, thethree input signals are input to an AND block A660. If the values of allthree inputs signals, are TRUE (i.e., ‘1’), the AND block A660 willproduce a TRUE (i.e., ‘1’) value for the combined trip signal indicatinga successful signal acquisition. If the values of any of the threeinputs is FALSE (i.e., ‘0’), the AND block A560 will produce a FALSE(i.e., ‘0’) value for the combined trip signal indicating that a signalhas not been successfully acquired and a search for a valid receivedsignal will continue.

Each transmitted message includes a fixed symbol as part of the leaderportion of the message. Since the fixed symbol causes no inter-symboldistortion, the receiver may use a trip correlator with a rectangular(unity) windowing function to maximize the correlation sum. Theresulting trip correlation function therefore achieves maximumsensitivity at the algorithmic point in time when it is most desirable(i.e., signal acquisition).

Following trip, a second correlation function is used for datademodulation. The data demodulation correlators use an alternatewindowing function to minimally impact sensitivity while providing someprotection to inter-symbol modulation distortion. In one embodiment ofthe present invention, this “dual-mode” QAM correlation is used tomaximize acquisition sensitivity while still providing inter-symbolinterference protection

FIG. 41 illustrates a dual-mode correlator according to one embodimentof the present invention. As shown in FIG. 41, the trip correlator A715receives the modulation domain signal of the FM detector A710 andproduces an output used by the trip detection A720 algorithm of thesystem. A separate data correlator A725 also receives the modulationdomain signal of the FM detector A710 and produces an output used by thedata demodulator A730. The QAM trip correlator A715 continues to receivethe modulation domain signal until a trip is detected in block A720.Once a trip event is detected, the data correlator A725 receives themodulation domain signal and delivers the modulation domain signal tothe data demodulator block A730. Neither the QAM trip correlator nor thedata demodulation QAM correlator have a DC (i.e., no error frequencycomponent) response. This has the very desirable benefit of completelyremoving the mean from the modulation domain signal (which relates tosystem frequency offset error) which alleviates the need for a meanremoval function in subsequent processing.

Correlator DC Removal For Frequency Error Tolerance

FIG. 42 illustrates the coefficients used in the trip QAM correlator andthe data decision QAM correlator for completely removing all DC responsewhile demodulating the QAM symbol constellation.

In one embodiment of the present invention a computationally efficientmethod is used in an incoherent FM receiver that tolerates frequencyerror between a transmitter and a receiver. The method includes twoelements. The first element is a frequency detector that is linear overthe range of twice the peak signal deviation plus twice the peakfrequency error. This frequency detector is described elsewhere in thisdisclosure. The second element follows the frequency detector. It is aQAM sub-demodulator that does not respond to DC. The QAM sub-demodulatorfirst stage consists of in-phase (I) and quadrature-phase (Q)correlators.

Conventional QAM optimum correlators are used in the signal detectionand signal acquisition method of the receiver. The coefficients of thesecorrelators are illustrated in FIG. 42 as I F120 and Q F110. Thesecorrelators have no response to DC because the coefficients sum to zero.Modified QAM correlators are used in the signal demodulation method ofthe receiver. The coefficients of these correlators are illustrated inFIG. 42 as I F140 and Q F130. These correlators also have no response toDC because the coefficients sum to zero. The corresponding transmitterfilters the modulating symbol waveforms to constrain the transmittedsignal spectrum to satisfy channel bandwidth requirements. Thisintroduces intersymbol interference with standard QAM optimumcorrelators. The triangle taper eliminates the intersymbol interferenceat a minor sensitivity expense.

In one embodiment of the present invention, the correspondingtransmitter sends a constant symbol leader. Therefore, there is nointersymbol interference on the leader. In this embodiment, conventionalQAM optimum correlators are used in the signal detection and signalacquisition method of the receiver.

Of course, it is possible to use a single correlator to simplifyconstruction with the associated loss of sensitivity in trip. Many ofthe diagrams of this disclosure show a QAM correlator that can be eithera single correlator or the dual-mode correlator of the presentinvention.

The QAM trip correlator A215 shown in FIG. 40 represents one exemplaryembodiment of the trip algorithm according to the present invention. Theaverager block A220 averages the trip correlator outputs over 16successive symbols producing smoothed data which is input to theCartesian-to-polar converter A225. Extending the averager beyond 16 willimprove sensitivity, but for the purposes of the present invention, alength of 16 was sufficient to achieve the necessary system bit errorrate (BER).

In one embodiment of the present invention the comparison time intervalis extended to 3 successive symbols from the simple model discussedabove. Additional delay elements can be added to enhance false triprejection at a cost of time to trip, but the system of the presentinvention uses two delay elements to balance trip sensitivity with datademodulation sensitivity. Although the system of FIG. 40 implements atwo-tap delay line, any number of additional delay elements can beadded. Delay blocks A230 and A231 form a two-tap delay line that yieldstwo older magnitudes. The output of A225, r(n), is the input to delayelement A230. The output of A230, r(n−1), is the input of delay elementA231. The output of A231 is r(n−2). Also, blocks A240 and A241 form atwo-tap delay line that yields two older angles, Θ(n−1) and Θ(n−2). Theoutput magnitude and angle terms are fed as inputs into FIGS. 43, 44, 45and 46, which are continuations of the trip algorithm.

Once the receive trip condition has been satisfied, the receiverperforms an alignment step to synchronize the receiver to transmitsymbol frame. FIG. 47 illustrates the synchronization step of the signalacquisition algorithm according to one embodiment of the presentinvention. As shown in FIG. 47, a delay element A805 controls the[time/phase] delay of a received time domain signal. Location of thisdelay in the signal thread is unimportant but must be far enoughupstream of decimation to allow for fine resolution time stepssufficient to achieve symbol alignment. In one embodiment of the presentinvention this delay is located at the digitizer sample rate andachieves a maximum symbol alignment error of 1.77 degrees. The delayA805 is shown in this diagram prior to FM demodulation A810 to simplifythe diagram.

The FM demodulator block A810 translates the time domain signal into amodulation domain signal. A QAM Correlator A815 receives the modulationdomain signal and generates I and Q data signals that are input to anaverager A820 to produce averaged I and Q data signals. Note that theQAM Correlator A815 in this embodiment of the present invention employsthe principles of the dual-mode correlator previously discussed. Duringthe acquisition portion of the algorithm (including the synchronizationsteps that follow), the trip correlators are used. The averaged I and Qdata signals are input into a Cartesian-to-Polar converter A825 togenerate r (magnitude) and Θ (angle) values.

The transmitted signal contains a leader portion with a known repeatingpattern of symbols. The leader symbol of the transmitter being receivedby the receiver will produce a constant Θ value from block A825. Thisvalue is the error (or difference) of the desired signal with respect tothe random initial phase of the receiver correlator. Since the receiverknows the desired angle associated with the leader symbol, a correctioncan be made by adjusting a phase delay element A805. The adjustment ismade by comparing the received Θ term during the leader against a storeddesired symbol Θ A830. Symbol alignment A840 compares the stored value(Θ) A830 with the detected angle value (Θ) A825. The difference betweenthe detected angle value A825 and the desired angle value A830 can betranslated to a radian shift term for use in the delay element A805 tosynchronize the incoming signal into the FM demodulator A810. Thesynchronization occurs only once upon valid signal detection delineatingacquisition and data demodulation.

The initial data decision threshold values are also calculated duringthe leader and prior to data demodulation. The calculated magnitude (r)A825 is used to set the data decision threshold values. Once trip hasbeen determined, the calculated r A825 is multiplied at a multiplierA855 by a threshold array A850 to create an array of data decisionthresholds that is used by the data demodulator A860. The array ofdecision thresholds A850 is predetermined based on the QAM spatialseparation created by the frequency tone separation of the FSK signal.In one embodiment of the present invention a 7 level FSK signal iscoupled to a 16QAM correlator to create a uniform constellation of 16symbols, each symbol encoding 4 bits of data.

FIG. 48 illustrates the derivation of the threshold array constants usedin one embodiment of the present invention. The symbol allocated for useas the leader symbol has the largest magnitude possible in theconstellation. In one embodiment of the present invention the “A” symbolis used. The constellation space is uniformly distributed such that eachpoint is 2 units of magnitude separated from its nearest neighbor A1010.A ratiotic set of threshold values is created using the received “A”symbol magnitude represented by the vector from the origin of the IQplane to symbol “A.” A single data decision threshold array can beapplied to either the I projection of the symbol vector or the Qprojection due to uniform placement of the constellation space. It canbe seen in the constellation of A1010 that the ideal slicer value forseparation of symbol A to symbol F or B as projected on the I axis wouldbe 2 (equidistant from the I axis intercept). One embodiment of thepresent invention dynamically calculates the “2” scaled to the actualreceived A symbol magnitude.

An ideal symbol A would have a magnitude of 3*SQRT(2) with an idealslicer value of 2. A constant scaling factor of SQRT(2)/3 is held instorage and applied to the measured magnitude of the leader symbol A,the product of which produces the slicer value for differentiating therightmost symbols of the constellation (A, E, 6 and 2) from theright-center constellation points (B, F, 7 and 3). Also, due to uniformconstellation space, the left-half plane slicer values are created bynegating the right-half plane values which were calculated (i.e., theconstellation is symmetrical about both the I and Q axis). Also, thecenter slicer values remain at zero due to the DC mean removal in thecorrelators.

The imaginary slicer values are identical to the real values, also dueto the uniform constellation as shown in A1020 and A1030 of FIG. 48.Changes in transmitter frequency deviation are translated to symbolmagnitude deviations that are compensated for by this algorithm.Accordingly, fundamentally low cost transmitters with imprecisefrequency deviation control may be used.

The process described above can be extended to larger QAM constellationsto create different threshold arrays.

Returning to FIG. 47, the threshold array is implemented as a either alookup table or constant values to ratiotically create a data slicerarray. The product term A855 uses the received magnitude of the tripsymbol to set the slicer array values. Uncertain deviation inherent inthe transmitted signal is therefore compensated for by the ratiotictreatment of the decision slicer values. This enables fundamentally lowcost methods for control of the frequency deviation on transmitters.

Once the data slicer array terms are calculated from the product A855,and the symbol alignment term has achieved synchronization in A805, thereceiver transitions from acquire mode to data demodulate mode. Anappropriate time delay prior to data interpretation is necessary toaccommodate filter settling. This time delay is dependent on filterimplementation and known in the art and, therefore, not shown in thefigures.

The dual-mode QAM correlators shift to the appropriate coefficients andthe data demodulator A860 uses the array of data decision thresholdsfrom A855 to translate I and Q data outputs from A815 to generate thesymbol value A870.

In one embodiment of the present invention, a decision feedback,adaptive algorithm is used for maintaining optimal data decisionthreshold (data slicer) values as shown in FIG. 49. The data decodeprocess according to one embodiment of the present invention is improvedby continually updating the data slicers nominated at signal detection(trip) throughout the message decode.

As shown in FIG. 49, the process begins at step A905 where it isdetermined if the adaptive slicer is enabled following the tripalgorithm. If the adaptive slicer is not active (i.e., “No” at stepA905), the decoded symbol is output at step A910 without upgrading thedata decision threshold array (slicer values). If the adaptive slicer isactive (i.e., “Yes” at step A905), the delay counter is tested at stepA915. The delay counter has the function of holding off the adaptivefunction until such a time as the signal reaches steady state followingtrip and the associated synchronization. The delay shift ofsynchronization inserts a perturbation into the system that takes a timeinterval to settle out. The delay counter A920 holds off the adaptiveslicer to perform this function.

Once the delay element has timed out (i.e., “Yes” at step A915), theadaptive slicer algorithm is enabled at step A925 where the symbol iscalculated using the most recently calculated slicer value from initialtrip or last symbol calculation. If the calculated symbol matches the“leader” symbol (i.e., “Yes” at step A925), the algorithm flow proceedsto step A945, which recalculates the appropriate slicer values based onthe latest available magnitude values scaled by the threshold array atstep A955 as disclosed previously in block A850 of FIG. 47. The processthen proceeds to step A965 where the slicer values are stored andsubsequently used during the next symbol data decode task. If thecalculated symbol does not match the “leader” symbol (i.e., “No” at stepA935), the receiver assumes the message portion of the received signalhas begun and the adaptive algorithm terminates at step A935. The lastcalculated slicer values are used for the remaining symbols throughoutthe message.

In one embodiment of the present invention, the adaptive data slicer isimplemented to compensate for low cost methods of deviation stability inthe transmitter. If the transmitter deviation stability drifts over theduration of the leader, this algorithm effectively tracks out the slowdrift component to improve BER for the message reception. In oneembodiment of the present invention, the transmitters exhibit adeviation drift that stabilizes in the leader portion of the message. Assuch, the adaptive equalization algorithm discussed above are sufficientto set slicer values by the start of the data portion of the message.

Alternatively, if the deviation is continuously changing, the adaptivealgorithm can be modified to continually adapt throughout the messagereception. In this mode, the function performed at step A925 is replacedby a normalization function to scale the magnitude of the receivedsymbol to that of the “leader” symbol prior to completing the functionpreviously discussed. All symbols are processed at step A945 throughoutthe decoded message. All constellation points are used to create newdecision thresholds for subsequent symbols.

The signal acquisition and adaptive slicer determination processdiscussed above can be used in all of the receiver architectures of thepresent invention. The detection algorithm can be enhanced to resolveeven wider frequency uncertainty than the FM demodulator can accommodateby using a plurality of channel filters and voting mechanism forselecting an optimal data path as described below.

Multi-channel Trip Technique

FIG. 50 illustrates a multi-channel trip algorithm according to oneembodiment of the present invention. As shown in FIG. 50, algorithmbegins with a channel filter bank D110. The channel filter bank D110accepts sequential inputs in the time domain and produces a plurality ofchannelized output data corresponding to frequency. In one embodiment awindowed-FFT is utilized to accomplish this, but any band-limitingfilter bank meeting the design criteria will suffice. The output of thechannel filter bank D110 is provided to element D115 which performs acomplex frequency downconversion to each channel output of the filterbank D110 from its respective center frequency to DC. The magnitude ofeach channel is then determined at element D120 (in another embodiment,magnitude squared may be used for ease of implementation in a DSP).Element D125 selects the index (channel number) of the channel havingthe largest magnitude as output from element D120. Element D130 averagestwo or more of the selected channel indices received from element D125and averages then decimates providing for an output rate that is L timesslower than the input rate. The averaged, selected channel indices arethen rounded to the nearest channel index by element D135. This roundedchannel index is used as an identifier to the channel containing thedesired signal.

In one embodiment of the present invention, the probability of selectingthe best channel for signal acquisition is increased by using therounded channel index, as well as its two neighboring indices, toidentify the channels selected for FM demodulation as seen in elementD140. Element D145 is a FM demodulator used to transform the incomingfrequency-domain channel data into the modulation domain. The leaderportion of the transmitted signal contains a repetition of a symbolpattern. In the modulation domain, the mean of the frequency deviationof this repeated symbol pattern is equal to zero when centered perfectlyin a channel. When the mean of the frequency deviation is not equal tozero in the modulation domain, then the frequency difference between thechannel center frequency and the down-converted symbol center frequencyis proportional to the mean of the frequency deviation. Elements D150and D155 are used to average and decimate the modulation domain data bytwo or more. The output of element D155 is the mean of the frequencydeviation corresponding to the selected input channels. The channelhaving a mean of the frequency deviation nearest to zero represents thechannel which is centered in frequency closest to the down-convertedsymbol center frequency, thus, being the best channel for further signalacquisition as seen in elements D160 and D165.

It may be possible to increase the probability of selecting the bestchannel for signal acquisition, if the rounded channel index is chosenalong with more than two neighboring channel indices to identify thechannels selected for FM demodulation and subsequent processing aspreviously described. There is a point of diminishing return when theinput bandwidth is less than the cumulative bandwidth of the channelsselected minus the amount of frequency overlap of these channels.

FIG. 51 shows a receiver incorporating the multi-channel trip accordingto one embodiment of the present invention. The function of thisreceiver is identical to that illustrated in FIG. 52 with the exceptionsthis receiver does not include elements B150 and B152, and this receiverreplaces elements B160 and B165 with element D260.

The above-described frequency uncertainty method works in systemapplications where the sample frequency F_(s) of the A/D D245, coupledwith the anti-alias filter D240 bandwidth, can be used to pass ananticipated signal that has frequency uncertainty well outside the FMdetector linear region. In many instances, it is desirable to use thereceiver architecture to scan a much wider bandwidth than iseconomically feasible by increasing the sample rate and anti-aliasbandwidth.

As the pass bandwidth of the system increases, the design requirementsof the digitizer D245 also increase to preserve two-toneinter-modulation performance as is known in the art. One embodiment ofthe present invention dwells momentarily at a specific frequency andthen slews the receiver local reference D225 to time division multiplexthe system over a wider bandwidth.

Systems typically require that the transmitter leader be long enough induration to allow for the receiver to sample all available frequenciesin order to achieve synchronization and data demodulation. Oneembodiment of the present invention dwells on up to 4 such frequencypassbands. Changes to transmitter leader length or acquisition time canchange the ratio as required.

Efficient signal acquisition may provide for pipelined processing ofdata while slewing frequency as depicted in FIG. 53. The localreference, element D225, should be tasked to the next frequency andallowed to settle while processing the data collected at the presentfrequency.

A method for extending the frequency resolving range of the receiver isto tune the local reference, element D225. By stepping the localreference in frequency steps, the subsequent digitization, signaldetection and signal acquisition may be performed at the bandwidth setby the IF filter D240, while maintaining the ability to search over awider range than the IF filter bandwidth.

Multi-Channel Receiver

The receiver architecture of the present invention lends itself tomultiple configurations described herein. Each configuration hasapplication in different communication scenarios.

FIG. 52 illustrates one embodiment of a receiver according to thepresent invention. In this embodiment, the receiver contains only thecomponents necessary to receive the M-ary FSK signal as described above.FIG. 52 illustrates a receiver architecture that may detect and decodedata from a compatible transmitter described above. As shown in FIG. 52,a typical heterodyne RF architecture is used to present the collectedsignal to a digitizer. Heterodyne downconversion techniques are wellknown in the art, and thus detail will not be provided herein ElementB110 represents an antenna or group of antennas used to collect radiofrequency (RF) energy for use in the receiver. The signal is bandlimited in element B115 prior to subsequent downconversion. The functionof B115 is to remove the alias image produced by the heterodynedownconversion process. Filter element B115 must remove energy outsidethe passband of interest to avoid undesired mixing against the frequencyreference.

Either high-side injection or low-side injection techniques can be usedin the downconversion process. High-side injection is when the localreference B125 is higher in frequency than the desired signal to bereceived at antenna B110. Low-side injection is when the local referenceB125 is lower in frequency than the desired signal to be received atantenna B110. In either case, the frequency separation between the localreference and the desired signal is equal to the desired IF centerfrequency.

Elements B120 and B135 represent distributed gain sufficient toestablish noise floor and signal sensitivity. Implementation for gain isnot described herein, as it is a well known practice in the art. ElementB125 and Element B130 frequency downconvert the desired RF signal to ananalog intermediate frequency (IF) whose output is filtered by elementB140 prior to digitization in element B145. Filter element B140 must besufficient to provide anti-alias protection in the digitizer B145,sufficiently rejecting images produced in the mixer B130 element causedby downconversion. The sample frequency applied to the digitizer B145 ischosen to enable the use of bandpass sampling techniques, thus creatinga digitized IF as an output of the digitizer B145. Bandpass sampling andIF center frequency selection are both well known techniques in the artand not described herein.

The resulting digitized IF signal will have sufficient bandwidth toallow for the desired signal, plus all frequency errors due to relativereference uncertainty, to pass without distortion observing at leastNyquist bandwidths of the sampler. More specifically, the desiredsignal, including all tolerable system errors, is passed withoutaliasing. All signal degradation contributors must be managed including,but not limited to, unacceptable quantization or noise floor degradationdue to noise aliasing.

Element B150 performs a final complex downconversion to center thedesired passband digital IF signal at baseband using a complexdownconversion input B152. Element B155 performs a time delay of theincoming digitized baseband samples. The time delay is adjusted by thereceive algorithm following trip (signal acquisition) to achieve datasymbol alignment (synchronization) to enable data demodulation.

Element B160 performs a channel filter function to noise limit thebaseband digitized IF signal to match the maximum desired signalbandwidth including anticipated frequency uncertainty typically set bytransmitter and receiver frequency reference errors. Typically, thesample rate of the digitized baseband IF signal is an order of magnitudehigher than the desired passband IF signal to be subsequently applied tothe FM demodulator B170. The output of the channel filter can bedecimated by block B165 to a minimum of the Nyquist bandwidth of thesignal plus frequency uncertainty to save computation in the FMdemodulator B170.

The FM demodulator B170 can be any of several methods to translate thetime domain sampled data into modulation domain referred to herein as afunction of omega or notated as ω(n). Block B170 depicts a multi-tapdifferentiator similar in structure as that taught by Frerking, M. E.,“Digital Signal Processing in Communication Systems,” ISBN0-442-01616-6, Kluwer Academic Publishers, Sixth Printing, 1994, pp.249-257, the entire contents of which is incorporated herein byreference. One embodiment of the present invention implements a 17 tapdifferentiator which has several features that enhance systemperformance. First, an odd number of taps provides for a center tap toextract the undifferentiated, delayed data path used in the demodulator.Second, extending the differentiator provides for a wider linear region,thus allowing larger frequency error in the signal. One embodiment ofthe present invention ensures that the largest frequency error presentin the signal creates less than 1 dB of compression in the FMdemodulator due to non-linearities in the multi-tap demodulator.

The output of the FM demodulator B170 is passed to a QAM demodulatorB175 which translates the modulation domain data into a quadratureconstellation in I and Q (real and imaginary). The modulation domainsignal contains a multi-level FSK with a DC offset that is proportionalto system frequency error. The QAM demodulator removes the inherent DCoffset that is present in the modulation domain signal. The techniquesand advantages of the QAM correlator were discussed above.

The output of the QAM correlator B175 is processed by a trip algorithmB180 to detect and acquire the signal during the leader of the messageusing the r-theta trip algorithm discussed above. Block B190 determinesproper symbol alignment timing and data decision threshold values basedon detected frequency deviations of the signal during the leader portionof the received message. Outputs of block B190 drive the delay elementB155 to achieve symbol synchronization and drives B185 to decode the QAMdata resulting in data bits for message decode in block B195.

The system of FIG. 52 provides for a receiver that can detect anddemodulate an ideal M-ary FSK/QAM transmitter. One embodiment of thepresent invention adds additional elements shown in FIG. 54 tocompensate for the non-ideal elements present in the optimally costadvantageous transmitter previously defined. Equalizer EQ1 B281 isinserted between the FM demodulator B170 and the trip and acquisitionQAM correlator B175. Equalizer EQ2 (B282) is inserted between the FMdemodulator B170 and the data QAM correlator B175. Both equalizerscompensate for distortions present in the transmitted signal due to costsaving techniques previously described herein. The equalizers eachaddress specific non-linearities present in the received signal causedby low cost methods for creating the signal. The transmitter PLL timeconstant which is roughly equal to the FSK symbol rate conserves batterylife but produces a bandpass modulation transfer function that steeplyrolls off below the symbol rate resulting in impact to systemsensitivity via intersymbol interference. The equalizer B282 compensatesfor the transmitter non-ideal elements recovering most of the signaldegradation loss created by the cost saving measures in the transmitter.

Similarly, equalizer B281 compensates for the IIR frequency responseproduced in the transmitter due to cost saving techniques previouslydisclosed. Equalizer B281 compensates for the non-ideal elements thatleft alone would dramatically impact intersymbol interference and reducesystem sensitivity. Both equalizers are unnecessary if the transmitterproduces a signal without the non-ideal elements discussed.

In another embodiment of the present invention, this one slightly morecomplex, the receiver architecture can simultaneously process multiplesignals by duplicating a portion of the receive thread. FIG. 55illustrates a typical implementation for this embodiment of the presentinvention when implementing a base station style of receiverarchitecture capable of performing cellular or redundant cellularfunction.

FIG. 55 includes a simplified RF downconversion block B315 whichincludes all the components previously discussed in the context of FIG.52 (B115, B120, B125, B130, B135, and B140). Block B315 can be anymethod for downconverting the signal from RF to IF as discussed inregard to FIG. 52 or otherwise. Figure elements B350, B352, B355, B360,B365, B370, B375, B380, B385, B390 and B395 form the same processingfunction as described in regard to FIG. 54. Similarly, the architecturepresented in FIG. 55 contains at least one more signal processing thread(notated as B351, B353, B356, B361, B366, B371, B376, B381, B386, B391and B396) denoting a separate demodulation capability. Together, thearchitecture provides the ability to simultaneously detect anddemodulate at least two signals separated by at least the bandwidth ofthe signal of interest plus allowable frequency error.

The number of simultaneously decodable channels is limited only by theNyquist bandwidth of the digitizers and the processing capability toperform the bandlimiting channel filters of each processing thread. Insome instances where the processing element includes a digitizingcapability, or the proximity of the digitizer to the processing elementis favorable, the digitizer may be duplicated per processing channel.FIG. 56 illustrates such an implementation. In this case, the processingelement B415 performs a function similar to that of B315 describedabove. Digitizing elements B445 and B446 are tightly coupled to theprocessing thread B450 and B451 respectively which each perform thefunctions described in regard to FIG. 52, elements B150, B152, B155,B160, B165, B170, B175, B180, B185, B190, B195.

Architectures

FIG. 57 illustrates a minimum differentiated system that provides thereceiver performance desirable as described earlier herein according toone embodiment of the present invention. Element C110 represents anantenna or group of antennas with the purpose of collecting RF energyfor use in the receiver. Block 112 collects elements C115, C120, C125,C130, C135 and C140, which together form a heterodyne downconversionarchitecture with the purpose of frequency translating the signal at RFto an IF signal prior to digitizing the signal in element C145. Thedigitized IF spectrum from element C145 is further downconverted tobaseband in mixer element C150 using the complex downconversion tonedenoted by C152. This baseband downconversion tone is typicallyperformed in hardware or firmware using digital signal processingtechniques known in the art.

The resulting baseband digitized signal is presented to delay elementC155 which will subsequently be used to achieve symbol alignment andsynchronization. The delay element C155 presents the digitized basebandIF signal to channel filter C160 whose function is to noise limit the IFsignal prior to FM demodulation in block C170. It is possible to performa decimation step prior to FM demodulation if the sample rate is higherthan necessary to satisfy the signal bandwidth plus frequencyuncertainty. In this case, a decimation in time step is desirable asdepicted in symbol C165. Preferably, the output effective sample ratefrom the decimation element is such that the Nyquist bandwidth of thesignal stream perfectly matches the desired signal bandwidth plusmaximum anticipated frequency uncertainty, such frequency uncertainty isset by errors in transmitter and receiver frequency references and anyother temporal error to frequency caused by system design issues such asDoppler.

One embodiment of the present invention implements the channel filterC160 using cascaded Kaiser filters. FIG. 58 depicts a set of replacementfilters for C160 including the Kaiser embodiment implementation C730.Any means of band limiting the signal prior to FM detection performs thesame function, but care must be taken to ensure that passband groupdelay does not cause undesired compression in the modulation domain. Forthis reason, flat group delay filters are preferred when possible, suchfilters to include FIR and Sinc as examples. IIR filters or filters withwide group delay swings may be used as long as they are approximatelylinear in the frequency band where the desired signal will operate.

The FM demodulator block, C170, performs a classical FM discriminatorfunction translating the time domain input signal to a modulation domainoutput. Such FM discriminators are known in the art and genericallyrepresented here. The resulting modulation domain signal is presented toa QAM correlator, C175, where the M-ary FSK modulation domain signal istranslated into a QAM constellation. The resulting signal is now incomplex (I, Q) space and can be translated into polar notation forsubsequent processing as r, Θ by the symbol alignment (synchronization)and data decision threshold determination functions.

Signal acquisition is performed in block C180 using the outputs of theQAM correlator using the r theta trip algorithm previously describedherein. Synchronization alignment terms and data decision thresholds areperformed in block C190 following successful signal acquisition in blockC180. Alignment terms are fed back to block C155. Data decisionthreshold terms are fed forward to the data decode block C185. Data isultimately decoded in block C185 presenting the data for messagecreation in block C195.

In another embodiment of the present invention, the heterodynedownconversion C112 function of FIG. 57 may be performed using an imagerejecting analog downconversion step illustrated in FIG. 59. An imagerejecting architecture allows for direct downconversion from RF atantenna element C210 using a pair of mixers C230, C231 that use areference frequency C225 shifted 90 degrees in phase C232 on one of themixers. The quadrature downconversion produces a single image centeredin the passband filters C240 and C241, which serve as anti-alias filtersfor the digitizing elements C245 and C246. The digitized complex IFsignal is presented to the channel filter and data demod block C260 thatperforms the functions previously disclosed to demodulate the M-aryFSK/QAM signal structure.

In yet another embodiment, the heterodyne downconversion C112 can bereplaced with a cascaded downconversion architecture using more than 1mixer as shown in FIG. 60. FIG. 60 illustrates two downconversion stagesbut standard practice dictates that any number or stages could beemployed paying special consideration to image frequencies produced toensure signal quality is not degraded. The system illustrated in FIG. 60shows the antenna C310 coupled to the first stage of the downconversionthread with the first heterodyne stage containing elements C315, C320,C330 and C325. The output of the first heterodyne stage feeds a secondstage with elements C340, C335, C337, C338, C342 and C344. The resultinganalog IF signal output is finally presented to digitizer C345 prior toprocessing by the channel filter and data demodulator block C350 whichperforms the functions previously disclosed to demodulate the M-aryFSK/QAM signal structure.

Another embodiment of the system shown in FIG. 57 uses a multi-tap FMdemodulator instead of the generic FM demodulator block. Extending thetime baseline in the demodulator and adding points can extend thefrequency error processing of the system. The specific implementationfor a multi-tap demodulator is known in the art and described in detailin Frerking as noted previously.

FIG. 61 illustrates a 5 tap differentiator style FM demodulator.Elements C410 and C420 depict the 5 tap delay lines, each holdingcomplex data and coefficients accordingly. Data is transitioned throughelement C410 while coefficients are static in C420. The centercoefficient holds a 0 value with a preferred length being odd. An oddlength differentiator provides for uniform group delay between theproduct-summation (C430, C440) and the d′ (d-prime) delayed path createdfrom the center tap of the data delay path C410. The 5 tapdifferentiator is a specific implementation of a known technique thatyields superior performance to a simple two tap model. Extending thetaps provides one tactic for improving frequency uncertainty range.

One embodiment of the present invention implements a 17 tapdifferentiator illustrated in FIG. 62. Similar to the 5 tapdifferentiator, data is passed through element C510 with a center tapproviding the delayed d′ data output. Coefficients are stored staticallyin element C520 with each new complex data input fed into the data delayelement C510 creating a product-sum in elements C530 and C540. Furthermodifications to the demodulator taps (e.g., to length N) may furtherimprove performance. One embodiment of the present invention implementsa QAM correlator which translates the M-ary FSK input to a quadratureconstellation for post processing. More specifically, the presentinvention translates 7 level FSK into a 16QAM constellation. The QAMcorrelator uses sequential time intervals of the 7 level FSK to encodethe QAM constellation. Still more specifically, the 7 specific frequencytones are patterned over 4 successive time intervals to encode 16discrete symbols as encoded in FIG. 63.

Each of the 7 frequency tones are represented in FIG. 63 as −3, −2, −1,0, 1, 2 or 3. Classical FSK rules apply such that each tone is uniformlyspaced to maximize receive sensitivity. 16 unique symbols are encoded bytransitioning four tones over time. Furthermore, each defined symboluses a set of tones which are centered about the center frequency, thusproducing an inherent whitening element useful in the transmittersection as described earlier herein. Each symbol, when translated to itsbinary equivalent, is so arranged as to provide for single bit errorcorrection due to a gray code allocation. A constellation error ineither the real or imaginary direction produces a single bit error whichis correctable using a simple cyclical redundant correction (CRC) codeembedded in the message.

PSK and ASK Modulation Method

In further embodiments of the present invention, rather than using anFSK modulation, phase shift keying (PSK) or amplitude shift keying (ASK)is used to modulate the data onto a carrier frequency. PSK and ASKmodulation techniques are well known to those of ordinary skill in thecommunications art. In these embodiments, the PSK or ASK modulator iscoupled to the QAM encoder and decoder to leverage the conceptsdescribed herein for other modulation techniques.

FIG. 64 illustrates that the FSK modulator of the present invention maybe replaced by, for example, a PSK modulator. FIG. 64 is a block diagramof a simple transmit device demonstrating an implementation of the PSKmodulator according to one embodiment. As shown in FIG. 64, anoscillator GG10 generates a carrier frequency used to transmit a signal.The oscillator GG10 represents a frequency generation capability at thetransmission frequency. In practice, the transmission frequency may becreated using any one of several known techniques including, but notlimited to, upconversion, synthesis, phase locked loops, or any numberof other techniques known in the art. For the purposes of thisdisclosure, the generation of the transmission frequency is simplifiedto demonstrate that the transmitter creates a frequency that issubsequently modulated to carry data. The PSK modulator of element GG12induces a discrete phase shift for the carrier frequency created by theoscillator GG10. The phase shift may be induced by a configurable timedelay or other mechanism to provide a fixed phase shift in frequencycontrollable by the L-State PSK controller GG11. The output of the PSKmodulator GG12 is provided to a bandpass filter GG14 to manage spectralgrowth due to phase shifting of the carrier signal. In practice, thisspectral re-growth may be managed directly by the PSK modulator GG12 bycontrolling the phase transition boundaries to limit phasediscontinuities. As would be understood by those of ordinary skill inthe communications art, the filtering process can be illustrated as anindependent element, such as bandpass filter GG14, while beingimplemented in another component of the transmit system. The bandpassfilter GG14 is optional, since it merely limits adjacent channelinterference. The gain element GG16 provides the power amplificationnecessary to transmit the signal. In practice, the gain may bedistributed across several elements in the device of the transmitthread. Finally, the signal leaves the transmitter through a radiatingelement GG18, such as an antenna.

Data is modulated on the carrier at the PSK modulator of GG12.Serialized data from the data source GG15 is encoded by the QAM encoderGG13. The QAM encoder GG13 breaks the serial data into symbols usingwhatever forward error correction or data packet method is desired. Inone embodiment of the present invention, the data is gray encoded into aQAM substructure. While in one embodiment of the present invention a 16QAM substructure is implemented, other constellations are also feasible.The 16 QAM encoder GG13 of the present invention breaks the serial datainto nibbles, gray encodes it onto the 16 QAM constellation, andpresents a word of data to the L-State PSK controller GG11.

The L-State PSK controller GG11 accepts the word of data from the QAMencoder GG13 and produces a signal that is usable by the PSK modulatorGG12. This signal may be analog or digital. The L-State PSK controllerGG11 drives the PSK modulator GG12 with a series of signals that shiftthe phase of the carrier frequency in a predetermined pattern based onthe QAM submodulation approach.

In one embodiment of the present invention, seven discrete phase shiftsin a unique set of patterns map to the 16 QAM states. Other phase shiftsand QAM submodulation constellation patterns are also possible.

FIG. 65 illustrates an exemplary receiver embodiment capable ofreceiving and demodulating a signal from a PSK/QAM transmitter. As withthe transmit device described above in the context of FIG. 64, thisfigure illustrates a generic architecture for the purposes ofdemonstrating an approach to demodulating a PSK signal that is coupledwith a QAM submodulation according to the present invention. The antennaGG20 collects the signal which is subsequently filtered by a passbandfilter GG21. The passband filter GG21 is an optional element, buttypical in receiver architectures, and primarily serves to band limitthe input signal for intermodulation protection in subsequent mixer oramplification elements. A receiver typically has at least onedownconversion element, as would be understood by those of ordinaryskill in the digital communication art. However, for brevity, thedownconversion has not been shown in FIG. 65, and is not discussedherein. Gain element GG22 represents a distributed gain for the receiverthat is sufficient to operate subsequent differentiation and datademodulation. A phase differentiator GG23 compares the received signalphase with that of a local reference GG28. The local reference may be aVCO or other frequency generation mechanism. In the example of FIG. 65,the local reference is a VCO that has two modes of operation. While thesignal is in acquisition mode, the track and hold element GG27 isconfigured to track. The error voltage created in the phasedifferentiator GG23 is fed back to the VCO to track out error and createa coherent receiver architecture. The diagram is not necessarilycomplete, as most feedback error correction techniques that are known inthe art employ filtering to limit oscillations. Once the signal isacquired, the track and hold element GG27 is configured to hold for theduration of the message reception. Changes in input phase due to datamodulation beyond the leader, appear on the error voltage node fordigitization by the analog to digital conversion (ADC) block GG24. Datademodulation in the QAM decoder GG25 operates on the sequence of phasesmeasured by the ADC element. The QAM decoder GG25 correlates the seriesof phases into a serial data stream GG26.

This feedback, closed loop architecture represents a coherent decodingmethod. In another embodiment, a non-coherent decoding method is used. Anon-coherent decoding method simply monitors the DC voltage on the errornode during the leader portion of the message and derives data relativeto that node voltage. Either method results in a method for demodulatingdata from the phase differentiator relative to the detected phase duringthe leader portion of a message.

The QAM decoder GG25 correlates the sequence of received phases(voltages) into symbols that reside on the QAM constellation. The outputof the QAM decode block is serial or symbol-grouped data.

In yet another variation of the present invention, amplitude shiftkeying is coupled to the QAM submodulation to modulate the data. FIGS.66 and 67 illustrate an exemplary transmitter and receiver that use thistechnique. Amplitude shift keying has some operational limitationscompared to FSK and PSK due to typical fading characteristics found inradio communication systems. However, for conducted communicationsystems, the ASK method may have utility.

FIG. 66 illustrates an exemplary transmit device configured to use anASK/QAM modulation technique. As shown in FIG. 66, a local oscillatorGG30 sets a transmission frequency and can be implemented by, forexample, a direct oscillator, or a combination of devices where a signalis generated and upconverted. A configurable attenuation element GG31 isdirect modulated to output the carrier frequency attenuated by data. Afilter GG32 is used to limit the spectral growth due to instantaneousamplitude shift keying. As in the PSK system described above, thisfiltering may be accomplished in several ways including direct slewlimiting in the attenuation element GG31. For illustration purposes, thefilter GG32 is included as a reminder that without filtering, someundesirable spectral growth may occur. Adjacent channel requirements ofthe system set the bandpass parameters for the filter. The gain elementGG33 represents the amplification necessary to transmit the signal atpower via the radiating element GG34. In practice this gain is oftendistributed throughout the transmit thread of components.

The data source GG37 provides data to be encoded by the QAM encoder GG36similar to the systems described above for FSK and PSK transmitters. Inthe ASK case, a series of L amplitudes are used to encode the QAMconstellation. In one embodiment of the present invention, 16 discreteconstellation points are mapped to 7 discrete amplitudes to create amodulation method where DC components on the transmitted AM waveform arezero. The QAM encoder GG36 performs this function for a given datasymbol to ensure the selected set of L state amplitudes have no DC mean.The L state controller GG35 merely steps out the sequence of amplitudescorresponding to the encoded symbol from the QAM encoder GG36.

FIG. 67 illustrates an exemplary receiver capable of demodulating asignal transmitted by an ASK/QAM transmitter. The antenna element GG40collects the signal and provides it to a filter GG41 to remove energyoutside the frequency of desired signal reception. The filter GG41 isused to limit intermodulation in subsequent downconversion oramplification mechanisms. A feedback error tracking system is used tomeasure the received amplitude of the signal during the leader and touse the measured amplitude as a reference during the data portion of themessage. This is accomplished using a sample and hold circuit GG47 thatmeasures the RMS voltage during the leader portion of the signal withthe switch closed, and holds that voltage during the data portion of themessage with the switch open. Alternatively, the DC mean of the signalcan be measured in the data, subtracting out the offset to correct in anopen loop system, thus reducing the need for sample and hold orcorrection circuits altogether.

A difference node GG43 subtracts the RMS voltage captured during theleader from the signal to produce a DC removed data signal to be sampledby the analog to digital converter (ADC) GG44.

An amplitude detector, for example the ADC GG44, measures the RMSamplitude of the signal and produces a voltage proportional to theamplitude of the signal. The ADC GG44 or other comparison mechanism canbe used to differentiate discrete states in amplitude. The ADC GG44measures the voltage changes which are relative to data modulation inthe signal and produces a series of amplitudes that are provided to theQAM decoder GG45 where the constellation symbols are decoded to serialdata GG46.

The ADC GG44 must have sufficient input range to eliminate thedifference block and sample and hold elements. The QAM decoder cansimply measure the ADC GG44 output during the leader and perform datademodulation relative to the measured ADC GG44 output.

As discussed above, the transmit and the receive hardware diagrams,FIGS. 64-67, are simplified diagrams that build on the principlesdescribed herein in the context of an FSK modulation technique. Thediagrams are illustrative of different techniques for coupling datamodulation to the QAM subcarrier. As in the case of the FSK/QAM method,the encoding and decoding mechanism can be accomplished without inducingDC into the modulation mechanism.

Frequency Uncertainty Resolution

In one embodiment of the present invention, a variety of compensationtechniques are provided to deal with frequency uncertainty in a transmitto receive communication system. The various techniques have previouslybeen disclosed herein, but not necessarily together. The following willserve to bring together the techniques and discuss their usefulness forreceiver systems with a variety of transmitter models.

When the transmitter and receiver components have tightly controlledfrequency references, frequency uncertainty is small and very slightaccommodation is necessary. As discussed above, one embodiment of thepresent invention includes a receive architecture whereby a simple, twotap FM differentiator is sufficient to pass the received signal,provided it was centered in the most linear region of the demodulator.

As the transmitter or receiver endpoints relax their frequency referencerequirements, it becomes necessary to extend the linear region of the FMdetector to accommodate the DC offset of in the modulation domain. Thesubsequent QAM demodulators have inherent DC removal in theimplementation. The system limitation remains the linearity of the FMdemodulator as it discriminates the signal with frequency uncertainty.The previous discussion disclosed methods implemented in variousembodiments of the present invention to reasonably extend the linearregion of the FM demodulator to accommodate frequency uncertainty.

As the frequency uncertainty between transmitter and receiver continuesto grow beyond the reasonable expansion of the FM demodulator,embodiments of the present invention use a method to process and aplurality of channelized filters with voting mechanisms to select anddirect a favorable channel to a FM demodulator for use. This techniqueworks well to balance the processing requirements of the FM processingthread with sample rate and passband components in the system.

As the frequency uncertainty between the transmitter and receivercontinues to grow beyond the desirable or possible plurality of filtersto resolve, it is possible to step the receiver local reference andblock translate a bandwidth of signal for processing in a deterministicsearch method as previously discussed. This type of frequency resolutionmethod has system impact which can be accommodated by several approachesincluding extending the transmitter leader as an example.

Together, the present invention provides for multiple methods ofresolving frequency uncertainty. The present invention implements all ofthese techniques toward different system implementations as dictated byapplication requirements.

The present invention is applicable to a variety of applications thathave heretofore been prohibitive or impractical due to associatedinfrastructure, wiring, and/or labor costs. For example, on a smallscale, the present invention enables many LAN-based applications. Theseapplications are typically installed within an environment that iseither owned or controlled by a user, such as, for example, a house, anoffice building, or a campus environment. These systems tend to supportapplications where the information gathered is time sensitive, and theinformation may be used for automating control. Applications include,but are not limited to, fire alarms, access control, security,HVAC/Energy management, lighting control, process control/industrialsensing, equipment monitoring (e.g., for vibration), remote moisturesensing/irrigation control, train health monitoring, local slow frameclosed circuit television, local asset tracking through an RFidentifier, halon dispensing, nurse call, tank leak detection, or reefercontainer temperature monitoring.

As another example, on a larger scale, the present invention enablesmany WAN-based applications. These applications are typicallygeographically distributed over an area that is not owned or controlledby any one user. These systems tend to support less time-criticalapplications where the information gathered is for management, billing,or some human action. Applications include, but are not limited to,utility meter reading, tank level monitoring (e.g., for re-supplyscheduling), termite detection, vending machine status monitoring,equipment service monitoring, vibration monitoring (e.g., for failureprediction), various GPS applications (e.g., location or routing),“home” arrest, package tracking, wide area asset tracking, intermodalfreight health, credit card verification, remote access HVAC control, orvarious quick response applications (e.g., chiller down).

The present invention may be deployed using different frequency bands asshown below in Table 1. Table 1 provides an exemplary list of severalfrequency bands at which the present invention may operate. However, thepresent invention is not limited to operation within the frequency bandslisted in Table 1.

TABLE 1 FROM TO 176 MHz 185 MHz 216 MHz 218 MHz 218 MHz 219 MHz 220 MHz222 MHz 223 MHz 235 MHz 868 MHz 872 MHz 902 MHz 928 MHz 955 MHz 960 MHz2.4 GHz 2.4835 GHz 5.725 GHz 5.85 GHz

FIG. 68 illustrates a computer system 1201 upon which an embodiment ofthe present invention may be implemented. The computer system 1201includes a bus 1202 or other communication mechanism for communicatinginformation, and a processor 1203 coupled with the bus 1202 forprocessing the information. The computer system 1201 also includes amain memory 1204, such as a random access memory (RAM) or other dynamicstorage device (e.g., dynamic RAM (DRAM), static RAM (SRAM), andsynchronous DRAM (SDRAM)), coupled to the bus 1202 for storinginformation and instructions to be executed by processor 1203. Inaddition, the main memory 1204 may be used for storing temporaryvariables or other intermediate information during the execution ofinstructions by the processor 1203. The computer system 1201 furtherincludes a read only memory (ROM) 1205 or other static storage device(e.g., programmable ROM (PROM), erasable PROM (EPROM), and electricallyerasable PROM (EEPROM)) coupled to the bus 1202 for storing staticinformation and instructions for the processor 1203.

The computer system 1201 also includes a disk controller 1206 coupled tothe bus 1202 to control one or more storage devices for storinginformation and instructions, such as a magnetic hard disk 1207, and aremovable media drive 1208 (e.g., floppy disk drive, read-only compactdisc drive, read/write compact disc drive, compact disc jukebox, tapedrive, and removable magneto-optical drive). The storage devices may beadded to the computer system 1201 using an appropriate device interface(e.g., small computer system interface (SCSI), integrated deviceelectronics (IDE), enhanced-IDE (E-IDE), direct memory access (DMA), orultra-DMA).

The computer system 1201 may also include special purpose logic devices(e.g., application specific integrated circuits (ASICs)) or configurablelogic devices (e.g., simple programmable logic devices (SPLDs), complexprogrammable logic devices (CPLDs), and field programmable gate arrays(FPGAs)).

The computer system 1201 may also include a display controller 1209coupled to the bus 1202 to control a display 1210, such as a cathode raytube (CRT), for displaying information to a computer user. The computersystem includes input devices, such as a keyboard 1211 and a pointingdevice 1212, for interacting with a computer user and providinginformation to the processor 1203. The pointing device 1212, forexample, may be a mouse, a trackball, or a pointing stick forcommunicating direction information and command selections to theprocessor 1203 and for controlling cursor movement on the display 1210.In addition, a printer may provide printed listings of data storedand/or generated by the computer system 1201.

The computer system 1201 performs a portion or all of the processingsteps of the invention in response to the processor 1203 executing oneor more sequences of one or more instructions contained in a memory,such as the main memory 1204. Such instructions may be read into themain memory 1204 from another computer readable medium, such as a harddisk 1207 or a removable media drive 1208. One or more processors in amulti-processing arrangement may also be employed to execute thesequences of instructions contained in main memory 1204. In alternativeembodiments, hard-wired circuitry may be used in place of or incombination with software instructions. Thus, embodiments are notlimited to any specific combination of hardware circuitry and software.

As stated above, the computer system 1201 includes at least one computerreadable medium or memory for holding instructions programmed accordingto the teachings of the invention and for containing data structures,tables, records, or other data described herein. Examples of computerreadable media are compact discs, hard disks, floppy disks, tape,magneto-optical disks, PROMs (EPROM, EEPROM, flash EPROM), DRAM, SRAM,SDRAM, or any other magnetic medium, compact discs (e.g., CD-ROM), orany other optical medium, punch cards, paper tape, or other physicalmedium with patterns of holes, a carrier wave (described below), or anyother medium from which a computer can read.

Stored on any one or on a combination of computer readable media, thepresent invention includes software for controlling the computer system1201, for driving a device or devices for implementing the invention,and for enabling the computer system 1201 to interact with a human user.Such software may include, but is not limited to, device drivers,operating systems, development tools, and applications software. Suchcomputer readable media further includes the computer program product ofthe present invention for performing all or a portion (if processing isdistributed) of the processing performed in implementing the invention.

The computer code devices of the present invention may be anyinterpretable or executable code mechanism, including but not limited toscripts, interpretable programs, dynamic link libraries (DLLs), Javaclasses, and complete executable programs. Moreover, parts of theprocessing of the present invention may be distributed for betterperformance, reliability, and/or cost.

The term “computer readable medium” as used herein refers to any mediumthat participates in providing instructions to the processor 1203 forexecution. A computer readable medium may take many forms, including butnot limited to, non-volatile media, volatile media, and transmissionmedia. Non-volatile media includes, for example, optical, magneticdisks, and magneto-optical disks, such as the hard disk 1207 or theremovable media drive 1208. Volatile media includes dynamic memory, suchas the main memory 1204. Transmission media includes coaxial cables,copper wire and fiber optics, including the wires that make up the bus1202. Transmission media also may also take the form of acoustic orlight waves, such as those generated during radio wave and infrared datacommunications.

Various forms of computer readable media may be involved in carrying outone or more sequences of one or more instructions to processor 1203 forexecution. For example, the instructions may initially be carried on amagnetic disk of a remote computer. The remote computer can load theinstructions for implementing all or a portion of the present inventionremotely into a dynamic memory and send the instructions over atelephone line using a modem. A modem local to the computer system 1201may receive the data on the telephone line and use an infraredtransmitter to convert the data to an infrared signal. An infrareddetector coupled to the bus 1202 can receive the data carried in theinfrared signal and place the data on the bus 1202. The bus 1202 carriesthe data to the main memory 1204, from which the processor 1203retrieves and executes the instructions. The instructions received bythe main memory 1204 may optionally be stored on storage device 1207 or1208 either before or after execution by processor 1203.

The computer system 1201 also includes a communication interface 1213coupled to the bus 1202. The communication interface 1213 provides atwo-way data communication coupling to a network link 1214 that isconnected to, for example, a local area network (LAN) 1215, or toanother communications network 1216 such as the Internet. For example,the communication interface 1213 may be a network interface card toattach to any packet switched LAN. As another example, the communicationinterface 1213 may be an asymmetrical digital subscriber line (ADSL)card, an integrated services digital network (ISDN) card or a modem toprovide a data communication connection to a corresponding type ofcommunications line. Wireless links may also be implemented. In any suchimplementation, the communication interface 1213 sends and receiveselectrical, electromagnetic or optical signals that carry digital datastreams representing various types of information.

The network link 1214 typically provides data communication through oneor more networks to other data devices. For example, the network link1214 may provide a connection to a another computer through a localnetwork 1215 (e.g., a LAN) or through equipment operated by a serviceprovider, which provides communication services through a communicationsnetwork 1216. In preferred embodiments, the local network 1214 and thecommunications network 1216 preferably use electrical, electromagnetic,or optical signals that carry digital data streams. The signals throughthe various networks and the signals on the network link 1214 andthrough the communication interface 1213, which carry the digital datato and from the computer system 1201, are exemplary forms of carrierwaves transporting the information. The computer system 1201 cantransmit and receive data, including program code, through thenetwork(s) 1215 and 1216, the network link 1214 and the communicationinterface 1213. Moreover, the network link 1214 may provide a connectionthrough a LAN 1215 to a mobile device 1217 such as a personal digitalassistant (PDA) laptop computer, or cellular telephone. The LANcommunications network 1215 and the communications network 1216 both useelectrical, electromagnetic or optical signals that carry digital datastreams. The signals through the various networks and the signals on thenetwork link 1214 and through the communication interface 1213, whichcarry the digital data to and from the system 1201, are exemplary formsof carrier waves transporting the information. The processor system 1201can transmit notifications and receive data, including program code,through the network(s), the network link 1214 and the communicationinterface 1213.

Obviously, numerous modifications and variations of the presentinvention are possible in light of the above teachings. It is thereforeto be understood that within the scope of the appended claims, theinvention may be practiced otherwise than as specifically describedherein.

What is claimed is:
 1. A phase-locked loop frequency synthesizer,comprising: an L-state pulse width modulator configured to receive areference frequency signal and at least one entry from a frequencytable, and to output at least one N/N+1 modulus signals corresponding tothe at least one entry from the frequency table; a divide by NN+1controllable modulus divider configured to receive the at least oneN/N+1 modulus signals and to divide the output frequency signal by theat least one N/N+1 modulus signals to generate a second referencefrequency signal; a phase frequency detector configured to receive thereference frequency signal and the second reference frequency signal andto generate an error signal; a filter network configured to receive theerror signal and to output a voltage; and a voltage controlledoscillator configured to receive the voltage and to generate the outputfrequency signal.
 2. The phase-locked loop frequency synthesizer ofclaim 1, wherein: the filter network comprises at least one notch filterbetween the phase frequency detector and the voltage controlledoscillator.
 3. The phase-locked loop frequency synthesizer of claim 2,wherein the at least one notch filter comprises a twin-T RC passivefilter.
 4. The phase-locked loop frequency synthesizer of claim 2,wherein the at least one notch filter comprises a twin-T RC activefilter.
 5. The phase-locked loop frequency synthesizer of claim 1,wherein: L is 68, N is 64, and the frequency table is configured tostore entries having values of 0 to 68.